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HSP50216
TM
Data Sheet
July 31, 2006
FN4557.4
Four-Channel Programmable Digital DownConverter
The HSP50216 Quad Programmable Digital DownConverter (QPDC) is designed for high dynamic range applications such as cellular basestations where multiple channel processing is required in a small physical space. The QPDC combines into a single package, a set of four channels which include: digital mixers, a quadrature carrier NCO, digital filters, a resampling filter, a Cartesian-to-polar coordinate converter and an AGC loop. The HSP50216 accepts four channels of 16-bit real digitized IF samples which are mixed with local quadrature sinusoids. Each channel carrier NCO frequency is set independently by the microprocessor. The output of the mixers are filtered with a CIC and FIR filters, with a variety of decimation options. Gain adjustment is provided on the filtered signal. The digital AGC provides a gain adjust range of up to 96dB with programmable thresholds and slew rates. A cartesian to polar coordinate converter provides magnitude and phase outputs. A frequency discriminator provides a frequency output via the FIR filter. Selectable outputs include I samples, Q samples, Magnitude, Phase, Frequency and AGC gain. The output resolution is selectable from 4-bit fixed point to 32-bit floating point. The maximum output bandwidth achievable using a single channel is at least 1MHz.
Features
* Up to 70MSPS Input * Four Independently Programmable Downconverter Channels in a single package * Four Parallel 16-Bit Inputs - Fixed or Floating Point Format * 32-Bit Programmable Carrier NCO with > 115dB SFDR * 110dB FIR Out of Band Attenuation * Decimation from 8 to >65536 * 24-bit Internal Data Path * Digital AGC with up to 96dB of Gain Range * Filter Functions - 1 to 5 Stage CIC Filter - Halfband Decimation and Interpolation FIR Filter - Programmable FIR Filter - Resampling FIR Filter * Cascadable Filtering for Additional Bandwidth * Four Independent Serial Outputs * 3.3V Operation * Pb-Free Plus Anneal Available (RoHS Compliant)
Applications
* Narrow-Band TDMA through IS-95 CDMA Digital Software Radio and Basestation Receivers
Ordering Information
PART NUMBER HSP50216KI PART MARKING HSP50216KI TEMP RANGE (C) PACKAGE PKG. NO V196.12x12 V196.12x12
* Wide-Band Applications: W-CDMA and UMTS Digital Software Radio and Basestation Receivers
-40 to 85 196 Ld BGA -40 to 85 196 Ld BGA (Pb-free)
HSP50216KIZ HSP50216KIZ (Note)
NOTE: Intersil Pb-free plus anneal products employ special Pb-free material sets; molding compounds/die attach materials and 100% matte tin plate termination finish, which are RoHS compliant and compatible with both SnPb and Pb-free soldering operations. Intersil Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020.
1
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. 1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc. Copyright Intersil Americas Inc. 2006. All Rights Reserved All other trademarks mentioned are the property of their respective owners.
HSP50216 Block Diagram
P TEST REGISTER
INPUT SELECT, FORMAT, DEMUX
LEVEL DETECTOR
SCLK A(15:0) ENIA I INPUT SELECT, FORMAT, DEMUX NCO / MIXER / CIC Q FIR FILTERS, AGC, CARTESIAN-TO-POLAR COORDINATE CONVERTER SYNCA SDIA SD2A CHANNEL 0
B(15:0) ENIB I INPUT SELECT, FORMAT, DEMUX NCO / MIXER / CIC Q FIR FILTERS, AGC, CARTESIAN-TO-POLAR COORDINATE CONVERTER SYNCB SDIB SD2B CHANNEL 1 C(15:0) ENIC I INPUT SELECT, FORMAT, DEMUX D(15:0) CHANNEL 2 ENID NCO / MIXER / CIC Q BUS ROUTING FIR FILTERS, AGC, CARTESIAN-TO-POLAR COORDINATE CONVERTER OUTPUT SELECT, FORMAT, SERIALIZE SYNCC SDIC SD2C
I INPUT SELECT, FORMAT, DEMUX NCO / MIXER / CIC Q
FIR FILTERS, AGC, CARTESIAN-TO-POLAR COORDINATE CONVERTER
SYNCD SDID SD2D
CHANNEL 3
CLK RESET SYNCI SYNCO P(15:0) ADD(2:0)
INTRPT P INTERFACE
RD or RD / WR
WR or DSTRB
P MODE
CE
2
HSP50216 Pinout
196 LEAD BGA TOP VIEW
1
2
3
4
5
6
7
8
9
10
11
12
13
14
A A5 B A3 C A1 D B15 E B13 F B11 G B9 H CLK J B7 K B5 L B3 M B1 N C15 P C13 C11 C9 C7 C5 C3 C1 ENIC D14 D12 D10 D8 D6 D4 C14 C10 C8 GND VCC GND VCC GND VCC D9 D7 D5 D2 B0 C12 C6 C4 C2 C0 D15 D13 D11 ENID D3 D1 D0 B2 ENIB CE RD WR VCC B4 P MODE P1 GND P0 GND B6 P3 VCC P2 VCC B8 P5 GND P4 GND GND P7 VCC P6 VCC B10 P9 GND P8 GND B12 P11 VCC P10 A0 B14 ADD2 RESET P13 P12 A2 A4 ENIA A12 A14 SD2A SD1B SD2B SD2C SD2D INTRPT P15 P14 A6 A8 A10 VCC GND VCC GND VCC GND SD1C SD1D ADD0 ADD1 A7 A9 A11 A13 A15 SD1A SYNCA SYNCB SCLK SYNCC SYNCD SYNCI SYNCO
POWER PIN GROUND PIN
SIGNAL PIN THERMAL BALL NC (NO CONNECTION)
3
HSP50216 Pin Descriptions
NAME POWER SUPPLY VCC GND INPUTS A(15:0) B(15:0) C(15:0) D15 D14 D13 D12 D11 D10 D9 D8 D7 D6 D5 D4 D3 D2 D1 D0 ENIA ENIB ENIC ENID CONTROL CLK SYNCI I I Input clock. All processing in the HSP50216 occurs on the rising edge of CLK. Synchronization Input Signal. Used to align the processing with an external event or with other HSP50216 devices. SYNCI can update the carrier NCO, reset decimation counters, restart the filter compute engine, and restart the output section among other functions. For most of the functional blocks, the response to SYNCI is programmable and can be enabled or disabled. Synchronization Output Signal. The processing of multiple HSP50216 devices can be synchronized by tying the SYNCO from one HSP50216 device (the master) to the SYNCI of all the HSP50216 devices (the master and slaves). Reset Signal. Active low. Asserting reset will halt all processing and set certain registers to default values. I I I I I I I I I I I I I I I I I I I I I I I Parallel Data Input bus A. Sampled on the rising edge of clock when ENIA is active (low). Parallel Data Input bus B. Sampled on the rising edge of clock when ENIB is active (low). Parallel Data Input bus C. Sampled on the rising edge of clock when ENIC is active (low). Parallel Data Input D15 or tuner channel A COF. Parallel Data Input D14 or tuner channel A COFSync. Parallel Data Input D13 or tuner channel A SOF. Parallel Data Input D12 or tuner channel A SOFSync. Parallel Data Input D11 or tuner channel B COF. Parallel Data Input D10 or tuner channel B COFSync. Parallel Data Input D9 or tuner channel B SOF. Parallel Data Input D8 or tuner channel B SOFSync. Parallel Data Input D7 or tuner channel C COF. Parallel Data Input D6 or tuner channel C COFSync. Parallel Data Input D5 or tuner channel C SOF. Parallel Data Input D4 or tuner channel C SOFSync. Parallel Data Input D3 or tuner channel D COF. Parallel Data Input D2 or tuner channel D COFSync. Parallel Data Input D1 or tuner channel D SOF. Parallel Data Input D0 or tuner channel D SOFSync. Input enable for Parallel Data Input bus A. Active low. This pin enables the input to the part in one of two modes, gated or interpolated. In gated mode, one sample is taken per CLK when ENI is asserted. Input enable for Parallel Data Input bus B. Active low. This pin enables the input to the part in one of two modes, gated or interpolated. In gated mode, one sample is taken per CLK when ENI is asserted. Input enable for Parallel Data Input bus C. Active low. This pin enables the input to the part in one of two modes, gated or interpolated. In gated mode, one sample is taken per CLK when ENI is asserted. Input enable for Parallel Data Input bus D. Active low. This pin enables the input to the part in one of two modes, gated or interpolated. In gated mode, one sample is taken per CLK when ENI is asserted. Positive Power Supply Voltage, 3.3V 0.15 Ground, 0V. TYPE DESCRIPTION
SYNCO
O
RESET
I
4
HSP50216 Pin Descriptions
NAME OUTPUTS SD1A O Serial Data Output 1A. A serial data stream output which can be programmed to consist of I1, Q1, I2, Q2, magnitude, phase, frequency (d/dt), AGC gain, and/or zeros. In addition, data outputs from Channels 0, 1, 2 and 3 can be multiplexed into a common serial output data stream. Information can be sequenced in a programmable order. See Serial Data Output Formatter Section. Serial Data Output 2A. This output is provided as an auxiliary output for Serial Data Output 1A to route data to a second destination or to output two words at a time for higher sample rates. SD2A has the same programmability as SD1A except that floating point format is not available. See Serial Data Output Formatter Section and Microprocessor Interface section. Serial Data Output 1B. See description for SD1A. Serial Data Output 2B. See description for SD2A. Serial Data Output 1C. See description for SD1A. Serial Data Output 2C. See description for SD2A. Serial Data Output 1D. See description for SD1A. Serial Data Output 2D. See description for SD2A. Serial Output Clock. Can be programmed to be at 1, 1/2, 1/4, 1/8, or 1/16 times the clock frequency. The polarity of SCLK is programmable. Serial Data Output 1A sync signal. This signal is used to indicate the start of a data word and/or frame of data. The polarity and position of SYNCA is programmable. Serial Data Output 1B sync signal. This signal is used to indicate the start of a data word and/or frame of data. The polarity and position of SYNCB is programmable. Serial Data Output 1C sync signal. This signal is used to indicate the start of a data word and/or frame of data. The polarity and position of SYNCC is programmable. Serial Data Output 1D sync signal. This signal is used to indicate the start of a data word and/or frame of data. The polarity and position of SYNCD is programmable. (Continued) DESCRIPTION
TYPE
SD2A
O
SD1B SD2B SD1C SD2C SD1D SD2D SCLK SYNCA SYNCB SYNCC SYNCD
O O O O O O O O O O O
MICROPROCESSOR INTERFACE P(15:0) ADD(2:0) WR or DSTRB I/O I I Microprocessor Interface Data bus. See "Microprocessor Interface" on page 29. P15 is the MSB. Microprocessor Interface Address bus. ADD2 is the MSB. See "Microprocessor Interface" on page 29. Note: ADD2 is not used but designated for future expansion. Microprocessor Interface Write or Data Strobe Signal. When the Microprocessor Interface Mode Control, P MODE, is a low data transfers (from either P(15:0) to the internal write holding register or from the internal write holding register to the target register specified) occur on the low to high transition of WR when CE is asserted (low). When the P MODE control is high this input functions as a data read/write strobe. In this mode with RD/WR low data transfers (from either P(15:0) to the internal write holding register or from the internal write holding register to the target register specified) occur on the low to high transition of Data Strobe. With RD/WR high the data from the address specified is placed on P(15:0) when Data Strobe is low. See "Microprocessor Interface" on page 29. Microprocessor Interface Read or Read/Write Signal. When the Microprocessor Interface Mode Control, P MODE, is a low the data from the address specified is placed on P(15:0) when RD is asserted (low) and CE is asserted (low). When the P MODE control is high this input functions as a Read/Write control input. Data is read from P(15:0) when high or written to the appropriate register when low. See "Microprocessor Interface" on page 29. Microprocessor Interface Mode Control. This pin is used to select the Read/Write mode for the Microprocessor Interface. Internally pulled down. See "Microprocessor Interface" on page 29. Microprocessor Interface Chip Select. Active low. This pin has the same timing as the address pins. Microprocessor Interrupt Signal. Asserted for a programmable number of clock cycles when new data is available on the selected Channel.
RD or RD/WR
I
P MODE CE INTRPT
I I O
5
HSP50216 Functional Description
The HSP50216 is a four channel digital receiver integrated circuit offering exceptional dynamic range and flexibility. Each of the four channels consists of a front-end NCO, digital mixer, and CIC-filter block and a back-end FIR, AGC and Cartesian to polar coordinate-conversion block. The parameters for the four channels are independently programmable. Four parallel data input busses (A(15:0), B(15:0), C(15:0) and D(15:0)) and four pairs of serial data outputs (SDxA, SDxB, SDxC, and SDxD; x = 1 or 2) are provided. Each input can be connected to any or all of the internal signal processing channels, Channels 0, 1, 2 and 3. The output of each channel can be routed to any of the serial outputs. Outputs from more than one channel can be multiplexed through a common output if the channels are synchronized. The four channels share a common input clock and a common serial output clock, but the output sample rates can be synchronous or asynchronous. Bus multiplexers between the front end and back end sections provide flexible routing between channels for cascading back-end filters or for routing one front end to multiple back ends for polyphase filtering or systolic arrays (to provide wider bandwidth filtering). A level detector is provided to monitor the signal level on any of the parallel data input busses, facilitating microprocessor control of gain blocks prior to an A/D converter. Each front end NCO/digital mixer/CIC filter section includes a quadrature numerically controlled oscillator (NCO), digital mixer, barrel shifter and a cascaded-integrator-comb filter (CIC). The NCO has a 32-bit frequency control word for 16.3mHz tuning resolution at an input sample rate of 70MSPS. The SFDR of the NCO is >115dB. The barrel shifter provides a gain of between 2-45 and 2-14 to prevent overflow in the CIC. The CIC filter order is programmable between 1 and 5 and the CIC decimation factor can be programmed from 4 to 512 for 5th order, 2048 for 4th order, 32768 for 3rd order, or 65536 for 1st or 2nd order filters. Each channel back end section includes an FIR processing block, an AGC and a cartesian-to-polar coordinate converter. The FIR processing block is a flexible filter compute engine that can compute a single FIR or a set of cascaded decimating filters. A single filter in a chain can have up to 256 taps and the total number of taps in a set of filters can be up to 384 provided that the decimation is sufficient. The HSP50216 calculates 2 taps per clock (on each channel) for symmetric filters, generally making decimation the limiting factor for the number of taps available. The filter compute engine supports a variety of filter types including decimation, interpolation and resampling filters. The coefficients for the programmable digital filters are 22 bits wide. Coefficients are provided in ROM for several halfband filter responses and for a resampler. The AGC section can provide up to 96dB of either fixed or automatic gain control. For automatic gain control, two settling modes and two sets of loop gains are provided. Separate attack and decay slew rates are provided for each loop gain. Programmable limits allow the user to select a gain range less than 96dB. The outputs of the cartesian-to-polar coordinate conversion block, used by the AGC loop, are also provided as outputs to the user for AM and FM demodulation. The HSP50216 supports both fixed and floating point parallel data input modes. The floating point modes support gain ranging A/D converters. Gated, interpolated and multiplexed data input modes are supported. The serial data output word width for each data type can be programmed to one of ten output bit widths from 4-bit fixed point through 32bit IEEE 754 floating point. The HSP50216 is programmed through a 16-bit microprocessor interface. The output data can also be read via the microprocessor interface for all channels that are synchronized. The HSP50216 is specified to operate to a maximum clock rate of 70MSPS over the industrial temperature range (-40oC to 85oC). The power supply voltage range is 3.3V 0.15V. The I/Os are not 5V tolerant.
6
HSP50216 Input Select/Format Block
TEST ENI SELECT (IWA *000 - 12 or GWA F804 - 12) P TEST REGISTER (GWA F807 - 15:0) TESTENBIT (IWA *000 - 11 or GWA F804 - 11) TESTENSTRB (GWA F808) A(15:0) ENIA B(15:0) MUX ENIB C(15:0) ENIC D(15:0) ENID NOTE: ENI* SIGNALS ARE ACTIVE HIGH (INVERTED AT THE I/O PAD) EXTERNAL DATA INPUT SELECT (IWA *000 - 14:13 or GWA F804 - 14:13) INPUT ENABLE HOLD OFF (ENABLED BY SYNCI) (GWA F802 - 30) DE-MULTIPLEX CONTROL (0-7) (IWA *000 - 6:4 or GWA F8O4 - 6:4) INTERPOLATED/GATED MODE (IWA *000 - 3 or GWA F804 - 3) MUX EXTERNAL/TEST SELECT (IWA *000 - 15 or GWA F804 - 15) OFFSET BINARY OR TWO's COMPLEMENT (IWA *000 - 10 or GWA F804 - 10) 11/3, 12/3, FIXED POINT 13/3, 14/2 OR (IWA *000 - 8:7 FLOATING POINT or GWA F804 - 8:7) (IWA *000 - 9 or GWA F804 - 9)
15:0
TESTEN
15:0 FORMAT MUX
MUX
FLOATING POINT TO FIXED POINT
R E G
15:0 DATA TO NCO / MIXER OR LEVEL DETECTOR
15:0
EN
PROGRAMMABLE DELAY
ENI DATA SAMPLE ENABLE
PN ENABLE PN (IWA *000 - 0) CARRIER OFFSET FREQUENCY (COF)
PN TO CARRIER NCO/MIXER SOF TO RESAMPLER NCO SOF SYNC TO RESAMPLER NCO ENABLE SOF (IWA *000 - 1)
COF TO CARRIER NCO/MIXER COF SYNC TO CARRIER NCO/MIXER ENABLE COF (1WA *000 - 2)
RESAMPLER OFFSET FREQUENCY (SOF) SOF SYNC
COF SYNC
Each front end block and the level detector block contains an input select/format block. A functional block diagram is provided in the above figure. The input source can be any of the four parallel input busses (See Microprocessor Interface section, Table 3, "CHANNEL INPUT SELECT/FORMAT REGISTER (IWA = *000h)," on page 32 or a test register loaded via the processor bus (see Microprocessor Interface section, Table 42, "mP/TEST INPUT BUS REGISTER (GWA = F807h)," on page 45). The input to the part can operate in a gated or interpolated mode. Each input channel has an input enable (ENIx, x = A, B, C or D). In the gated mode, one input sample is processed per clock that the ENIx signal is asserted (low). Processing is disabled when ENIx is high. The ENIx signal is pipelined through the part to minimize delay (latency). In the interpolated mode, the input is zeroed when the ENIx signal is high, but processing inside the part continues. This mode
inserts zeros between the data samples, interpolating the input data stream up to the clock rate. On reset, the part is set to gated mode and the input enables are disabled. The inputs are enabled by the first SYNCI signal. The input section can select one channel from a multiplexed data stream of up to 8 channels. The input enable is delayed by 0 to 7 clock cycles to enable a selection register. The register following the selection register is enabled by the non-delayed input enable to realign the processing of the channels. The one-clock-wide input enable must align with the data for the first channel. The desired channel is then selected by programming the delay. A delay of zero selects the first channel, a delay of 1 selects the second, etc.
7
HSP50216
The parallel input busses are 16 bits wide. The input format may be twos complement or offset binary format. A floating point mode is also supported. The floating point modes and the mapping of the parallel 16-bit input format is discussed below. bit 15 (MSB): 20, bit 14: 2-1, bit 13: 2-2, ..., bit 0: 2-15. For floating point modes, the least significant 2 or 3 bits are used as exponent bits (See Floating Point Input Mode Bit Mapping Tables). The difference between the four floating point modes with three exponent bits is where the exponent saturates.
Floating Point Input Mode Bit Mapping
The input bit weighting for fixed point inputs on busses A, B, C, and D is:
Floating Point Input Mode Bit Mapping Tables
A(15:0), B(15:0), C(15:0) or D(15:0):
15 20 14 2-1 13 2-2 12 2-3 11 2-4 10 2-5 9 2-6 8 2-7 7 2-8 6 2-9 5 2-10 4 2-11 3 2-12 2 2-13/(exp2) 1 (exp1) 0 (exp0)
11-BIT MODE: 11 to 13-BIT MANTISSA, 3-BIT EXPONENT, 30dB EXPONENT RANGE EXPONENT 000 001 010 011 100 101 (Note 1) NOTES: 1. Or 110 or 111, the exponent input saturates at 10. 2. "Xnn" = input A, B, C, or D bit nn. 12-BIT MODE: 12 to 13-BIT MANTISSA, 3-BIT EXPONENT, 24dB EXPONENT RANGE EXPONENT 000 001 010 011 100 (Note 3) NOTE: 3. Or 101, 110, or 111, the exponent input saturates at 100. 13-BIT MODE: 13-BIT MANTISSA, 3-BIT EXPONENT, 18dB EXPONENT RANGE EXPONENT 000 001 010 011 (Note 4) NOTE: 4. Or 100, 101, 110, or 111, the exponent input saturates at 011. 0 6 12 18 GAIN (dB) X15 X15 X15 X15 X15 X15 X15 X14 X15 X15 X14 X13 X15 X14 X13 X12 PIN BIT WEIGHTING TO 16-BIT INPUT MAPPING X14 X13 X12 X11 X13 X12 X11 X10 X12 X11 X10 X9 X11 X10 X9 X8 X10 X9 X8 X7 X9 X8 X7 X6 X8 X7 X6 X5 X7 X6 X5 X4 X6 X5 X4 X3 X5 X4 X3 0 X4 X3 0 0 X3 0 0 0 0 6 12 18 24 GAIN (dB) X15 X15 X15 X15 X15 X15 X15 X15 X15 X14 X15 X15 X15 X14 X13 X15 X15 X14 X13 X12 PIN BIT WEIGHTING TO 16-BIT INPUT MAPPING X15 X14 X13 X12 X11 X14 X13 X12 X11 X10 X13 X12 X11 X10 X9 X12 X11 X10 X9 X8 X11 X10 X9 X8 X7 X10 X9 X8 X7 X6 X9 X8 X7 X6 X5 X8 X7 X6 X5 X4 X7 X6 X5 X4 X3 X6 X5 X4 X3 0 X5 X4 X3 0 0 X4 X3 0 0 0 0 6 12 18 24 30 GAIN (dB) X15 X15 X15 X15 X15 X15 X15 X15 X15 X15 X15 X14 X15 X15 X15 X15 X14 X13 X15 X15 X15 X14 X13 X12 PIN BIT WEIGHTING TO 16-BIT INPUT MAPPING X15 X15 X14 X13 X12 X11 X15 X14 X13 X12 X11 X10 X14 X13 X12 X11 X10 X9 X13 X12 X11 X10 X9 X8 X12 X11 X10 X9 X8 X7 X11 X10 X9 X8 X7 X6 X10 X9 X8 X7 X6 X5 X9 X8 X7 X6 X5 X4 X8 X7 X6 X5 X4 X3 X7 X6 X5 X4 X3 0 X6 X5 X4 X3 0 0 X5 X4 X3 0 0 0
8
HSP50216
14-BIT MODE: 14-BIT MANTISSA, 2-BIT EXPONENT, 12dB EXPONENT RANGE EXPONENT 00 01 10 (Note 5) NOTE: 5. Or 11, the exponent input saturates at 10. 0 6 12 GAIN (dB) X15 X15 X15 X15 X15 X14 X15 X14 X13 X14 X13 X12 PIN BIT WEIGHTING TO 16-BIT INPUT MAPPING X13 X12 X11 X12 X11 X10 X11 X10 X9 X10 X9 X8 X9 X8 X7 X8 X7 X6 X7 X6 X5 X6 X5 X4 X5 X4 X3 X4 X3 X2 X3 X2 0 X2 0 0
Level Detector
An input level detector is provided to monitor the signal level on any of the input busses. Which input bus, the input format, and the level detection type are programmable (see Microprocessor Interface section, Table 39, "INPUT LEVEL DETECTOR SOURCE SELECT/FORMAT REGISTER (GWA = F804h)," on page 44, Table 40, "INPUT LEVEL DETECTOR CONFIGURATION REGISTER (GWA = F805h)," on page 45 and Table 41, "INPUT LEVEL DETECTOR START STROBE REGISTER (GWA = F806h)," on page 45). This signal level represents the wideband signal from the A/D and is useful for controlling gain / attenuation blocks ahead of the converter. The supported monitoring modes are: integrated magnitude (like the HSP50214 without the threshold), leaky integration (Yn = Xn x A + Yn-1 x (1-A)) where A = 1, 2-8, 2-12, or 2-16 (see GWA = F805h), and peak detection. The measurement interval can be programmed from 2 to 65537 samples (or continuous for the leaky integrator and peak detect cases). The output is 32 bits and is read via the P interface.
After the mixers, a PN (pseudonoise) signal can be added to the data. This feature is provided for test and to digitally reduce the input sensitivity and adjust the receiver range (sensitivity). The effect is the same as increasing the noise figure of the receiver, reducing its sensitivity and overall dynamic range. For testing, the PN generator provides a wideband signal which may be used to verify the frequency response of a filter. The one bit PN data is scaled by a 16-bit programmable scale factor. The overall range for the PN is 0 to 1/4 full scale (see IWA = *001h). A gain of 0 disables the PN input. The PN value is formed as
PN Value
2-3 2-4 . SSS X X
. . . . . . . . . . .
2-17 2-18 X X
XXXXXXXXXXXX
where S is the PN generator output bit (treated as a sign bit) and the 16 X's refer to the PN Gain Register IWA = *001h. The minimum, non-zero, PN value is 2-18 of full scale (-108dBFS) on each axis (-105dBFS total). For an input noise level of -75dBFS, this allows the SNR to be decreased in steps of 1/8dB or less. The I and Q PN codes are offset in time to decorrelate them. The PN code is selected and enabled in the test control register (F800h). The PN is added to the signal after the mix with the three sign bits aligned with the most significant three bits of the signal, so the maximum level is 12dBFS and the minimum, non-zero level is -108dBFS. The PN code can be 215-1, 223-1 or 215-1 * 223-1.
NCO/Mixer
After the input select/format section, the samples are multiplied by quadrature sine wave samples from the carrier NCO. The NCO has a 32-bit frequency control, providing sub-hertz resolution at the maximum clock rate. The quadrature sinusoids have exceptional purity. The purity of the NCO should not be the determining factor for the receiver dynamic range performance. The phase quantization to the sine/cosine generator is 24 bits and the amplitude quantization is 19 bits. The carrier NCO center frequency is loaded via the P bus. The center frequency control is double buffered - the input is loaded into a center frequency holding register via the P interface. The data is then transferred from the holding register to the active register by a write to a address IWA *006h or by a SYNCI signal, if loading via SYNCI is enabled. To synchronize multiple channels, the carrier NCO phase accumulator feedback can be zeroed on loading to restart all of the NCOs at the same phase. A serial offset frequency input is also available for each channel through the D(15:0) parallel data input bus (if that bus is not needed for data input). This is legacy support for HSP50210 type tracking signals. See IWA=*000 and *004 for carrier offset frequency parameters.
9
HSP50216
CIC Filter
Next, the signal is filtered by a cascaded integrator/comb (CIC) filter. A CIC filter is an efficient architecture for decimation filtering. The power or magnitude squared frequency response of the CIC filter is given by:
2N sin ( Mf ) ----------------------- P(f) = sin f --- R
The CIC filter order is programmable from 0 to 5. The minimum decimation is 4. If the order is set to 0, there must be at least 4 clocks between samples or the decimation counter must be set to 4 to chose every 4th sample. The integrator bit widths are 69, 62, 53, 44, and 34 for the 1st through 5th stages, respectively, while the comb bit widths are all 32. The integrators are sized for decimation factors of up to 512 with 5 stages, 2048 with 4 stages, 32768 with 3 stages, and 65536 with 1 or 2 stages. Higher decimations in the CIC should be avoided as they will cause integrator overflow. In the HSP50216, the integrators are slightly oversized to reduce the quantization noise at each stage.
where M = Number of delays (1 for the HSP50216) N = Number of stages and R = Decimation factor. The passband frequency response for 1st (N=1) though 5th (N=5) order CIC filters is plotted in Figure 8. The frequency axis is normalized to fS/R, making fS/R = 1 the CIC output sample rate. Figure 10 shows the frequency response for a 5th order filter but extends the frequency axis to fS/R = 3 (3 times the CIC output sample rate) to show alias rejection for the out of band signals. Figure 9 uses information from Figure 10 to provide the amplitude of the first (strongest) alias as a function of the signal frequency or bandwidth from DC. For example, with a 5th order CIC and fS/R = 0.125 (signal frequency is 1/8 the CIC output rate) Figure 9 shows a first alias level of about -87 dB. Figure 9 is also listed in table form in Table 47.
10
HSP50216 Backend Data Routing
MAG: I AGC LOOP FILTER PATH 0 MUX GAIN dphi/dt: Q
I1 Q1
(4:0) M U X FROM CIC FILTER COMPUTE ENGINE
PATH 1 FIFO/ TIMER AGC MULT
x1, x2 x4, x8 CART TO POLAR SHIFT d/dt
MAG
M U X
PHASE
PATH 2
I2 Q2 EXT AGC GAIN DESTINATION BIT MAP (BITS 28:18 OF FIR INSTRUCTIONS BIT FIELD) 28 27 26 25 24 23 22 21 20 19 18 AGC LOOP GAIN SELECT (PATH 01 ONLY) UPDATE AGC LOOP (PATH 01 ONLY) PATH 00 - - IMMEDIATE FILTER PROCESSOR FEEDBACK PATH 01 - - FIFO/AGC PATH 10 - - DIRECT OUT/CASCADE PATH 11 - - BOTH 00 AND 10 PATHS (FOR TEST) STROBE OUTPUT SECTION (START SERIAL OUTPUT WITH THIS SAMPLE) FEED MAG/PHASE BACK TO FILTER PROCESSOR FILTER PROCESSOR SEQUENCE STEP NUMBER
28 27 26, 25
24 23 22:18
A CIC filter has a gain of RN, where R is the decimation factor and N is the number of stages. Because the CIC filter gain can become very large with decimation, an attenuator is provided ahead of the CIC to prevent overflow. The 24 bits of sample data are placed on the low 24 bits of a 69 bit bus (width of the first CIC integrator) for a gain of 2-45. A 32 bit barrel shifter then provides a gain of 20 to 231 inclusive before passing the data onto the CIC. The overall gain in the pre-CIC attenuator can therefore be programmed to be any one of 32 values from 2-45 to 2-14, inclusive (see IWA=*004, bits 18:14). This shift factor is adjusted to keep the total barrel shifter and CIC filter between 0.5 and 1.0. The equation which should be used to compute the necessary shift factor is: Shift Factor = 45 - Ceiling(log2(RN)).
NOTE: With a CIC order of zero, the CIC shifter does not have sufficient range to route more than 10 bits to the back end since the maximum gain is 2-14 (the least significant 14 bits are lost).
Back End Section
One back-end processing section is provided per channel. Each back end section consists of a filter compute engine, a FIFO/timer for evenly spacing samples (important when implementing interpolation filters and resamplers), an AGC and a cartesian-to-polar coordinate conversion block. A block diagram showing the major functional blocks and data routing is shown above. The data input to the back end section is through the filter compute engine. There are two other inputs to the filter compute engine, they are a data recirculation path for cascading filters and a magnitude and d/dt feedback path for AM and FM filtering. There are seven outputs from each back end processing section. These are I and Q directly out of the filter compute engine (I2, Q2), I and Q passed through the FIFO and AGC multipliers (I1, Q1), magnitude (MAG), phase (or d/dt), and the AGC gain control value (GAIN). The I2/Q2 outputs are used when cascading back end stages. The routing of signals within the back end processing section is controlled by the filter compute engine. The routing information is embedded in the instruction bit fields used to define the digital filter being implemented in the filter compute engine.
11
HSP50216 Filter Compute Engine
R E G 9..-31 0..-23 L I M I T R E G M U X
1..-25 WITH RND 0..-23 IQ M U X I Q RAM 384 WORDS I Q S W A P A B A L U 1..-23
DOWN SHIFT 0, 1, 2 PLACES
S H F T R E G S H F T
R/d/dt
0..-23 INMUX (1:0)
S W A P
A A L U 0..-21 ENFB, RNDSEL (2:0)
RAMR/Wb ADDRA (8:0) ADDRB (8:0)
B
R E G
L I M I T
R E G
M U X
COEF
RAMAEN
RAMBEN
QFUNCT
SHIFT (1:0)
REGEN4
COEF (21:0), SHIFT (1:0)
NOTE: PIPELINE DELAYS OMITTED FOR CLARITY
The filter compute engine is a dual multiply-accumulator (MAC) data path with a microcoded FIR sequencer. The filter compute engine can implement a single FIR or a set of filters. For example, the filter chain could include two halfband filters, a shaping (matched) filter and a resampling filter, all with different decimations. The following filter types are currently supported by the architecture and microcode: * Even symmetric with even # of taps decimation filters * Even symmetric with odd # of taps decimation filters (including HBFs) * Odd symmetric with even # of taps decimation filters * Odd symmetric with odd # of taps decimation filters * Asymmetric decimation filters * Complex filters * Interpolation filters (up to interpolate by 4) * Interpolation halfband filters * Resampling filters (under resampler NCO control) * Fixed resampling ratio filter (within the available number of coefficients) * Quadrature to real filtering (w/ fs/4 up conversion) The input to the filter compute engine comes from one of three sources - a CIC filter output (which can also be another backend section), the output of the filter compute engine (fed back to the input) or the magnitude and d/dt fed back from the cartesian-to-polar coordinate converter.
The number and size of the filters in the chain is limited by the number of clock cycles available (determined by the decimation) and by the data and coefficient RAM/ROM resources. The data RAM is 384 words (I/Q pairs) deep. The data addressing is modulo in power-of-2 blocks, so the maximum filter size is 256. The block size and the block starting memory address for each filter is programmable so that the available memory can be used efficiently. The coefficient RAM is 192 words deep. It is half the size of the data memory because filter coefficients are typically symmetric. ROMs are provided with halfband filter coefficients, resampling filter coefficients, and constants. The filter compute engine exploits symmetry where possible so that each MAC can compute two filter taps per clock, by doing a pre-add before multiplying. In the case of halfband filters, the zero-valued coefficients are skipped for extra efficiency. There is an overhead of one clock cycle per input sample for each filter in the chain (for writing the data into the data RAM) and (except in special cases) a two clock cycle overhead for the entire chain for program flow control instructions. The output of the filter compute engine is routed through a FIFO in the main output path. The FIFO is provided to more evenly space the FIR outputs when they are produced in bursts (as when computing resampling or interpolation filters). The FIFO is four samples deep. The FIFO is loaded by the output of the filter when that path is selected. It is unloaded by a counter. The spacing of the output samples is specified in clock periods. The spacing can be set from 1 (fall through) to 4096 samples
12
OUTSEL
ENLIMIT
IFUNCT
IQSWAP
ENHR1
ENHR2
HSP50216
(approximately the spacing for a 16KSPS output sample rate when using 65MSPS clock) using IWA = *00Ah bits 11:0. The number and order of the filtering in the filter chain is defined by a FIR control program. The FIR control program is a sequence of up to 32 instruction words. Each instruction word can be a filter or program flow instruction. The filter instruction defines a FIR in the chain, specifying the type of FIR, number of taps, decimation, memory allocation, etc. For program flow, a wait for input sample(s) instruction, a loop counter load, and several jumps (conditional and unconditional) are provided. The HSP50216 evaluation board includes software for automatically generating FIR control programs for most filter requirements. Examples of programs FIR control programs are given below. The simplest filter program computes a single filter. It has three instructions (see Sample Filter #1 Program Instructions below):
SAMPLE FILTER #1 PROGRAM STEP 0 1 INSTRUCTION Wait for enough input samples (equal to the decimation factor) FIR Type = even symmetric 95 taps Decimate by 2 Compute one output Decrement wait counter Memory block size 128 Memory block start at 64, Coefficient block start at 64 Step size 1 Output to AGC Jump, Unconditional, to step 0 3
rate to the FIR from the CIC filter would be 2.5MSPS. The impulse response length would be 38 sec (95 taps at 0.4s/tap). Each additional filter added to the signal processing chain requires one instruction step. As an example of this, a typical filter chain might consist of two decimate-by-2 halfband filters being followed by a shaping filter with the final filter being a resampling filter. The program for this case might be (see Sample Filter Program #2 Program Instructions below):
SAMPLE FILTER #2 PROGRAM STEP 0 1 INSTRUCTION Wait for enough input samples (usually equal to the total decimation -- 8 in this case) FIR Type = even symmetry 15 taps Halfband Decimate by 2 Compute four outputs Memory block size 32 Memory block start at 0 Coefficient block start at 13 Output to step 2 Decrement wait count FIR Type = even symmetry 23 taps Halfband Decimate by 2 Compute two outputs Memory block size 32 Memory block start at 32 Coefficient block start at 24 Output to step 3 FIR Type = even symmetry 95 taps Decimate by 2 Compute one output Memory block size 128 Memory block start at 64 Coefficient block start at 64 Step size 1 Output to step 4 FIR Type = resampler Increment NCO 6 taps Compute one output Memory block size 8 Memory block starts at 192 Coefficient block start at 512 Step size 32 Output to AGC Jump, Unconditional, to 0
2
2
The parameters of the FIR (including type, number of taps, decimation and memory usage) are specified in the bit fields of the step 1 instruction word. To change the filtering the only other change needed is the number of samples in the wait threshold register (IWA = *00C, bits 9:0). The filter in this example requires 52 clock cycles to compute, allocated as follows:
SAMPLE FILTER #1 CLOCK CYCLES CALCULATION CLOCK CYCLES 48 2 2 52
4 FUNCTION PERFORMED Clocks for FIR computation (two taps/clock due to symmetry) Clocks for writing the input data into the data RAMs (Decimate by 2 requires 2 inputs per output) Clocks for the program flow instructions (wait and jump) Total 5
Using a 65MSPS clock, the output sample rate could be as high as 65MSPS / 52 clocks = 1.25MSPS. The input sample
13
HSP50216
Sample filter #2 requires: * 32 + 32 + 128 + 8 = 200 data RAM locations * (95+1)/2=48 coefficient RAM location (resampler and HBF coefficients are in ROM). The number of clock cycles required to compute an output for Sample filter #2 is calculated as follows:
SAMPLE FILTER #2 CLOCK CYCLES CALCULATION CLOCK CYCLES 20 8 14 4 48 2 6 1 1 1 105 FUNCTION PERFORMED Halfband 1 compute clocks (5 per compute x 4 computes) Halfband 1 input sample writes (8 input samples) Halfband 2 compute clocks (7 per compute x 2 computes) Halfband 2 input sample writes (4 input samples) 95 tap symmetric FIR, 2 clocks per tap FIR input sample writes (2 input samples) resampler (6 taps, nonsymmetric) Resampler input sample write (1 input samples) Jump instruction Wait instruction Clock cycles per output CLOCK CYCLES 6 2 15 1 1 1 26 FUNCTION PERFORMED 19 tap halfband, one output halfband input writes (2 input samples) 30 tap symmetric FIR, 2 taps per clock 1 FIR input write 1 wait 1 jump Clock cycles per output 3 2 STEP 1 SAMPLE FILTER #3 PROGRAM (Continued) INSTRUCTION FIR Type = even symmetry 19 taps Halfband Decimate by 2 Compute one output Memory block size 32 Memory block start at 0 Coefficient block start at 18 Output to step 2 Reset wait count FIR Type = even symmetry 30 taps Decimate by 1 Compute one output Memory block size 64 Memory block start at 32 Coefficient block start at 64 Step size 1 Output to AGC Jump, Unconditional, to 0
The number of clock cycles required to compute an output for Sample filter #3 is calculated as follows:
SAMPLE FILTER #3 CLOCK CYCLES CALCULATION
Total decimation is 8, so the input sample rate for the FIR chain (CIC output rate) could be up to: fCLK/(ceil(105/8)) = fCLK/14. With a 65MHz clock, this would support a maximum input sample rate to the FIR processor of 4.6MHz and an output sample rate up to 0.580MHz. The shaping filter impulse response length would be: (95 x 2)/580,000 = 82s. The maximum output sample rate is dependent on the length and number of FIRs and their decimation factors. Illustrating this concept with Filter Example #3, a higher speed filter chain might be comprised of one 19 tap decimate-by-2 halfband filter followed by a 30 tap shaping FIR filter with no decimation. The program for this example could be:
SAMPLE FILTER #3 PROGRAM STEP 0 INSTRUCTION Wait for enough input samples (2 in this case)
For Filter Example #3 and a 65MSPS input, the maximum FIR input rate would be 65MSPS / ceil(26 / 2) = 5MSPS giving a decimate-by-2 output sample rate of 2.5MSPS. At 70MSPS, the FIR could have up to 34 taps with the same output rate. Channels 0, 1, 2 and 3 can be combined in a polyphase structure for increased bandwidth or improved filtering. Filter Example #4 will be used to demonstrate this capability. Symbol rate of 4.096 MSym. The desired output sample rate is 8.192MSPS. Arrange the four back end sections as four filters operating on the same CIC output at a rate of 65.536MHz/4=16.384MHz, where the factor of 4 is the CIC decimation we have chosen. Each channel computes the same sequence, offset by one output sample from the previous sample (see IWA = *00Bh). Each channel decimates down to 2.048M and then the
14
HSP50216
channels are multiplexed together in the output formatter to get the desired 8.192MSPS. The input sample rate to the final filter of each channel must meet Nyquist requirements for the final output to assure that no information is lost due to aliasing.
SAMPLE FILTER #4 PROGRAM STEP 0 1 INSTRUCTION Wait for enough input samples (8 in this case) FIR type = even symmetry 44 taps decimate by 8 compute one output memory block size 64 memory block start at 0 coefficient block start at 64 step size 1 output to AGC offset memory read pointers by 0, -2, -4, -6 Jump, Unconditional, to 0
consisting of condition code selects, FIR parameters and data routing controls. Not all of the instruction word bits are used for all instruction types. The actual sequencer instruction is only 9 bits. The rest of the bits are used for filter parameters or for the loop counter preload. Each sequence step is loaded in four 32-bit writes. The mapping of the bit fields for the instruction types is shown in the instruction bit field table that follows. These FIR instruction words can be generated using software tools provided with the HSP50216 evaluation board. When the filter is reset, the instruction pointer is set to 31 (the last instruction step). The read and write pointers are initialized on reset, so a reset must be done when the channel is initialized or restarted. A fixed offset can be added to the starting read address of one of the filters in the program. This function is provided to offset the data reads of the filters in a polyphase filter bank -all filters in the bank will write the same data to the same RAM location. To offset the computations the RAM read address is offset. See IWA = *00Bh for details. The instruction word bits (127:0) are assigned to memory words as follows: 31:0 to destination C C C C 0 0 0 1 0 x x x x x 0 0 63:32 to destination C C C C 0 0 0 1 0 x x x x x 0 1 95:64 to destination C C C C 0 0 0 1 0 x x x x x 1 0 127:96 to destination C C C C 0 0 0 1 0 x x x x x 1 1 where CCCC is the channel number and xxxxx is the instruction sequence step number (0 - 31 decimal). Note the PHold bit in the filter compute engine control register (IWA = *00Ah) must be set for the microprocessor to read from or write to the instruction or coefficient RAMs.
2
The number of FIR taps available for these requirements is calculated as follows: 65536/2048 = 32 clocks minus (8 writes + 1 wait + 1 jump = 10 clocks) = 22 clocks Therefore, the number of taps available is: 22 x 2 = 44 taps. Multiplexing the four outputs gives a final output sample rate of 8.192MSPS. The impulse response is 44 taps at 16.384M or 22 output samples (11 symbols at 4.096M). The AGC loop filter output of channel 4 can be routed to control the forward AGC gain control of all four channels. This assures that the gains of the four back end sections are the same. The gain error, however, is only computed from every fourth output sample. The back end processing sections of two or more HSP50216s can be combined using the same polyphase approach, but the AGC gain from one part cannot be shared with another part (except via the P interface), so polyphase filter using multiple parts would typically usually use a fixed gain. The filter sequencer is programmed via an instruction RAM and several control registers. These are described below.
Instruction RAMs
The filter compute engine is controlled by a simple sequencer supporting up to 32 steps. Each step can be a filter or one of four sequence flow instructions - wait, jump (conditional or unconditional), load loop counter, or NOP. There are 128 bits per instruction word with each word
15
HSP50216 Filter Sequencer
FIR# - WRITE DESTINATION NEW DATA, FIR # RESET SYNC INSTRUCTION RAM, SEQUENCER FIR# - COMPUTE ALIAS MASK READ POINTER REG FILE WRITE POINTER REG FILE FIR OUTPUT DESTINATION THRESHOLD DECREMENT 1 DECREMENT 2 WAIT COUNTER DATA ADDRESS STEP SIZE COMPUTE TO COMPUTE FIR TYPE
START ADDRESS DATA PATH DATA PATH CONTROL CONTROL SIGNALS ROM
NUMBER OF OUTPUTS
TAPS/OUTPUT LOOP COUNTER READS/TAP LOOP COUNTER PRELOAD INSTR/TAP RAM ADDR BLOCK START RAM ADDR BLOCK SIZE RAM ADDR STEP SIZE 1 RAM ADDR STEP SIZE 2 FIR PARAMETER RAM ADDR BLOCK TO BLOCK STEP RAM COMPUTE COUNTERS
RAM ADDR GEN A
DATA RAM A READ/WRITE ADDRESS
RAM ADDR INITIAL OFFSET RAM ADDR OFFSET STEP RAM ADDR BLOCK TO BLOCK STEP RAM ADDR GEN B
DATA RAM B READ ADDRESS
ENABLE OFFSET
COEF ADDR BLOCK START COEF ADDR BLOCK SIZE COEF ADDR SIZE PER TAP ADDR STEP SIZE PER OUTPUT COEF ADDR GEN COEFFICIENT READ ADDRESS
RESAMPLER NCO
ADDRESS OFFSET
16
HSP50216 Instruction Bit Fields
INSTRUCTION BIT FIELDS BIT POSITIONS 8:0 FUNCTION Instruction Instruction Field Bit Mapping Bit Type WAIT FIR JUMP XXXX JJJJJ CCC 000 001 010 011 100 101 110 111 Start IncrRS 0 0 1 0 1 J = ignored. = jump destination (sequence step number). = condition code. = (waitcount threshold) -- See IWA = *00Ch, bits 9:0 for threshold details. = waitcount threshold -- See IWA = *00Ch, bits 9:0 for threshold details. = loop counter 0. = loop counter = 0. = RSCO Tab (RSCO - resampler NCO carry output). = RSCO. = sync (if enabled) or P controlled bit. = always. = load parameters and start filter computation, set to zero for no-ops, loop counter loads. = increment resampler during this filter. Increments on start or at each FIR output depending on Pcontrol bit. = decrement wait count on starting this instruction. = load loop counter with the data in the I(20:9) bit field. The start bit should not be set when this bit is set. = decrement loop counter on starting this instruction. = enable U/C counter with this FIR. This multiplies the data by 1, j, -1, -j. The multiplication factor changes each time the filter runs. X Start J X IncrRS J X J X J C LdLp C C DecrLp C C EnU/C C DecrSel DecrEn 8 7 6 5 4 3 2 1 0 DESCRIPTION
(NOPs and loading the loop counter are special cases of the FIR instruction).
DecrSel = selects between two decrement values for the wait counter. DecrEn LdLp DecrLp EnU/C
14:9
FIR Type
FIR Parameter Bit Fields 14:9 FIR type. 000000 000001 000010 000011 000100 000101 001000 001001 100000 NOTES: 1. Regular interpolation FIRs are successive runs of a FIR with no data address increment, but with coefficient start address increments. 2. Decimating HBFs are even symmetric, odd number of taps but with different data step sizes. 3. U/C FIR is a normal FIR with the U/C bit enabled. 4. Other codes may be added in the future. NOP. Decimating FIR, Even Symmetric, Even # Taps. Decimating FIR, Even Symmetric, Odd # Taps. Decimating FIR, Odd Symmetric, Even # Taps. Decimating FIR, Odd Symmetric, Odd # Taps. Decimating FIR, Asymmetric. Resampling FIR, Asymmetric. Interpolating HBF. Decimating FIR, Complex (Asymmetric).
17:15
Steps per FIR
Specifies the number of steps per FIR instruction sequence (load with value minus 1) (set to 0 for all FIR types except complex which is set to 1).
17
HSP50216
INSTRUCTION BIT FIELDS (Continued) BIT POSITIONS 28:18 FUNCTION Destination Destination Field Bit Mapping 28 27 26 AGCLFGN AGCLF Path1 25 Path0 DESCRIPTION 24 OS 23 FB 22 F4 21 F3 20 F2 19 F1 18 F0
AGCLFGNAGC loop gain select. Only applies to Path 1. Loop gain 0 or 1 if AGCLF bit is set. Set to 0 (1 is a test mode for future chips). AGCLF AGC loop filter enable. Only applies to Path 1. The AGC loop is updated with the magnitude of this sample (Path(1:0) = 01). 00Route output back to filter compute engine input to another FIR in the filter chain. 01Route output through the FIFO and AGC forward path to the cartesian-to-polar coordinate converter conversion and output (I1, Q1, magnitude, phase, gain) and also to route to a discriminator (i.e., d/dt FIR). 10Route output directly to the output, bypassing the FIFO and AGC (I2, Q2). This path also routes to next channel FIR input. OS Enable output strobe. Setting this bit generates a data ready signal when the data reaches the output section and starts the serial output sequence (paths 1, 2, 3). If OS is not set, there will be no output to the outside world from this channel, for that output calculation, but the data will be loaded into its output holding register (OS would not be set when routing the data to another back end when cascading channels). Feedback data path. When set, the magnitude and dphi/dt from the cartesian-to-polar coordinate converter block are routed to the filter compute engine input (magnitude goes to the I input and dphi/dt goes to the Q input). Provided for discriminator filtering. Filter select. For data recirculated to the input of the FIR processor by path 0 or from the cartesian to polar coordinate converter output, these bits tell which filter sequencer step gets it as an input. Round Select (Add rounding bit at specified location). 2-24, use this code when downshifting is not used. 2-23 2-22 2-21 2-20 2-19 2-18 no rounding.
Path(1:0) Back End Data Routing Path Selection.
FB
F(4:0)
31:29
Round Select
31:29 000 001 010 011 100 101 110 111
Provided for use with the coefficient down-shift bits. 41:32 44:42 Data Memory Block Start Data Memory Block Size Memory block base address, 0-1023, 0-383 are valid for the HSP50216. 44:42 0 1 2 3 4 5 6 7 Block Size. 8 16 32 64 128 256 512 1024
(modulo addressing is used). 52:45 Data Memory Block-to-Block Step 0-255, usually equal to the decimation factor for the FIR in this instruction.
18
HSP50216
INSTRUCTION BIT FIELDS (Continued) BIT POSITIONS 62:53 63 66:64 FUNCTION Coefficient Memory Block Start Reserved Coefficient Memory Block Size DESCRIPTION Memory base address of coefficients, 0-1023, 0-511 are valid on the HSP50216. Set to 0. 66:64 Memory Block Size 0 1 2 3 4 5 6 7 8 16 32 64 128 256 512 1024
(Modulo addressing can be used, but is usually not needed. If not needed this bit field can always be set to 7). 75:67 84:76 93:85 95:94 104:96 Number of FIR Outputs Read Address Pointer Step Initial Address Offset Reserved Memory Reads Per FIR Output Number of FIR outputs (range is 1 to 512, load w/ desired value minus 1). This is usually equal to the total decimation that follows the filter. Read address pointer step (for next run). This is usually equal to the filter decimation times the number of outputs from the instruction. Initial address offset (to ADDRB). This is the offset from the start address to other end of filter. For symmetric filters, usually equal to -1 x (number of taps -1). Set to 0 This is based on the number of taps (load with value below minus 1). Type Value Symmetric Symmetric Decimating HBF Asymmetric Complex Resampling Interpolating HBF 106:105 115:107 117:116 Clocks Per Memory Read Data Memory Step Size 1 Data Memory Step Size 2 even number of taps(taps/2) or floor((taps+1)/2). odd number of taps (taps+1)/2 or floor((taps+1)/2). (taps+5)/4. taps. taps. taps/phase (six taps per phase for the ROM'd coefficients provided). (taps+5)/4-1.
Set to 0 for all but complex FIR, which is set to 1. (ADDRA) Step size for all but the last tap computation of the FIR. Set to -2 for HBF, -1 otherwise. (ADDRA) Step size for last tap computation. Set to -1. 117:116 Step size 0 1 2 3 0 -1 -2 step size value.
119:118
Data Memory Address Offset Step
(ADDRB) Step size for opposite end of symmetric filter. Set to +2 for Decimating HBF, to +1 for others (the B data is not used for asymmetric, resampling, and complex filters).
19
HSP50216
INSTRUCTION BIT FIELDS (Continued) BIT POSITIONS 122:120 FUNCTION Coefficient Memory Step Size (ADDRC) Usually set to 1. 122:120 Step size 0 1 2 3 4 5 6 7 125:123 Coefficient Memory Block-to-Block Step 0 1 2 4 8 16 32 64 DESCRIPTION
(ADDRC) Usually set to 0. 125:123 Step size 0 1 2 3 4 5 6 7 0 1 2 4 8 16 32 64
127:126
Reserved
Set to 0
Basic Instruction Set Examples
1. Wait for number of input samples > threshold 127:9 = 0 8:0 = 001 0000,0000,0000,0001h 2. Jump unconditional 127:9 = 0 8:0 = 1JJJJJ111b example: jump to step 0= 0000,0000,0000,0107h 3. Jump RSCO (jump on resampler NCO carry output) 127:9 = 0 8:0 = 1JJJJJ101b example: jump RSCO, step 0= 0000,0000,0000,0105h 4. Jump RSCO (jump on no resampler NCO carry output) 127:9 = 0 8:0 = 1JJJJJ100b example: jump RSCO, step 0 = 0000,0000,0000,0104h 5. NOP single clock 127:9 = 0 8:0 = 010000000b NOP1 = 0000,0000,0000,0080h 6. Load Loop Counter 127:21 = 0 20:9 = Loop counter preload (tested against 0) 8:0 = 010000100b example: LdLpCntr 14 = 0000,0000,0000,1C84h
20
HSP50216
Single FIR Basic Program
This is the basic program for a single FIR. This program applies to decimation filters (including DECx1) that are symmetric or asymmetric (but not complex). The FIR output is routed through path A with the AGC enabled.
0 - WAIT FOR ENOUGH SAMPLES 0000 0000 0000 0000 1 - FIR 0000 00TT 0000 0000 0001 TTTT 1000 1011 0101 TTTD 0000 0000 1111 DDDD 0000 0000 1111 DDDD 0000 0FFF 100R 0000 1010 FFF0 RRRR 0000 0000 1100 RRRR 0111 0000 1000 127:96 95:64 63:32 31:0 015FF---h -----007h 08000A00h 0B00--C8h 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0001 127:96 95:64 64:32 31:0 00000000h 00000000h 00000000h 00000001h
2 - JUMP TO STEP 0 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0001 0000 0000 0000 0000 0000 0000 0000 0111 127:96 95:64 64:32 31:0 00000000h 00000000h 00000000h 00000107h
Four bit fields must be filled in: F - filter type (this example applies to types 1-5) D - decimation (also loaded into wait threshold) T - number of taps minus 1 R - clocks/calculation (=floor((taps+1)/2) for symmetric, = taps for asymmetric) The rest of the instruction RAM would typically be filled with NOP instructions: 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 0000 1000 0000 0000 0000 0000 127:96 95:64 64:32 31:0 00000000h 00000000h 00000000h 00000080h
Wait Preload Register
This register (IWA register *00Ch) holds the wait counter threshold and two wait counter decrement values. Each is 10 bits. The wait counter counts filter input samples until the count is greater than or equal to the threshold. The wait counter then asserts a flag to the filter compute engine. The wait counter threshold is typically set to the total number of input samples needed to generate a filter output. A "WAIT" instruction in the filter compute engine waits for the wait counter flag signal before proceeding. The filter compute engine would then compute all the filters needed to produce an output and then would jump back to the "WAIT" instruction. The wait counter is implemented with an accumulator. This allows the count to go beyond the threshold without losing the sample count. Two bits in the FIR instruction decrement the wait counter (subtract a value) and select the decrement value. The decrement value is typically the number of samples needed for an output (total decimation), though it can be a different value to ignore inputs and shift the timing. (The read pointer increment must be adjusted as well.) The filter compute engine sequencer does not count each input sample or track whether each filter is ready to run. Instead, the wait counter is used to determine whether there are enough input samples to compute all the filters in the chain and get an output sample from the entire filter chain. This adds some additional delay since intermediate results are not precalculated, but it simplifies the filter control. The number of samples needed is equal to the total decimation of the filter chain. For example, with two decimate-by-2 halfband filters and a decimate-by-2 shaping FIR, the total decimation would be 8 so 8 samples are needed to compute an output. HBF1 would compute four times to generate four inputs to HBF2. HBF2 would compute twice to generate the two samples that the shaping FIR needs to compute an output.
21
HSP50216 Resampler
The resampler is an NCO controlled polyphase filter that allows the output sample rate to have a non-integer relationship to the input sample rate. The filter engine can be viewed conceptually as a fixed interpolate-by-32 filter, followed by an NCO controlled decimator. The Resampler NCO is similar to the carrier NCO phase accumulator but does not include the SIN/COS section. It provides the resampler output pulse and associated phase information to logic that determines the nearest of the 32 available phase points for a given output sample. The center frequency (output sample rate) control is double buffered, i.e., the control word is written to one register via the microprocessor interface and then transferred to another (active) register on a write to the timing NCO center frequency update strobe location (IWA register *009h) or on a SYNCI (if enabled). As it is not possible to represent some frequencies exactly with an NCO and therefore, phase error accumulates eventually causing a bit slip, the phase accumulator length has been sized to where the error is insignificant. At a resampler input rate of 1MHz, half an LSB of error in loading the 56-bit accumulator is 7*10-12 degrees. After 1 year, the accumulated phase error is only 0.2*10-3 of a bit (< 1/10 of a degree). The NCO update by the filter compute engine is typically at the resampler's input rate, and is enabled by the IncrRS bit in the filter instruction word. The NCO then rolls over at a fraction of the resampler input rate. The output sample rate is (fIN/ 256)*N, where fIN is the resampler input rate and N is the phase accumulated per resampler input sample. N must be between 40000000000000h and FFFFFFFFFFFFFFh corresponding to decimations from 4 to (1 + 2-56), respectively. Generally, however, a range of 80000000000000h to FFFFFFFFFFFFFFh (providing decimation from 2 to (1 + 2-56), respectively) is sufficient for most applications since integer decimation can be done more efficiently in the preceding CIC and halfband filters. The resampler changes the sample rate by computing an output at each input which causes the NCO to roll over. If an output is to be computed, the nearest of the 32 available points from the polyphase structure is used. Because outputs are generated only on input samples which cause an NCO roll over, output samples will in general not be evenly spaced. The FIFO/TIMER block between the filter compute engine and the AGC is provided to improve output sample spacing for presentation to the serial data output formatter section (see IWA=*00Ah bits 11:0 description). If D/A converted directly, there would be artifacts from the uneven sample spacing, but if the samples are stored and reconstructed at the proper rate (the NCO rollover rate), the signal would have only the distortion produced by interpolation image leakage and the time quantization (phase jitter) due to the finite number of interpolation filter phases. The polyphase filter has 192 coefficients implemented as 32 phases, each of which having 6 taps (6 x 32 = 192). These coefficients are provided in Table 50. The stopband attenuation of the filter is greater than 60dB, as shown in Figures 13 - 15. The signal to total image power ratio is approximately 55dB, due to the aliasing of the interpolation images. If the output is at least 2x the baud rate, the 32 interpolation phases yield an effective sample rate of 64x the baud rate or approximately 1.5% (1/64 resampler input sample period) maximum timing error.
AGC
The AGC Section provides gain to small signals, after the large signals and out-of-band noise have been filtered out, to ensure that small signals have sufficient bit resolution in the output formatter. The AGC can also be used to manually set the gain. The AGC optimizes the bit resolution for a variety of input amplitude signal levels. The AGC loop automatically adds gain to bring small signals from the lower bits of the 24bit programmable FIR filter output into the range of 20-bit and shorter words in the output section. Without gain control, a signal at -72dBFS = 20log 10 (2 -12 ) at the input would have only 4 bits of resolution at the output if a 16 bit word length were to be used (12 bits less than the full scale 16 bits). The potential increase in the bit resolution due to processing gain of the filters can be lost without the use of the AGC. Figure 1 shows the Block Diagram for the AGC Section. The FIR filter data output is routed to the Cartesian to polar coordinate converter after passing through the AGC multipliers and shift registers. The magnitude output of the Cartesian to polar coordinate converter is routed through the AGC error detector, the AGC error scaler and into the AGC loop filter. This filtered error term is used to drive the AGC multiplier and shifters, completing the AGC control loop. The AGC multiplier / shifter portion of the AGC is identified in Figure 1. The gain control from the AGC loop filter is sampled when new data enters the multiplier / shifter. The limit detector detects overflow in the shifter or the multiplier and saturates the output of I and Q data paths independently. The shifter has a gain from 0 to 90.31dB in 6.021dB steps, where 90.31dB = 20log 10 (2 N ) when N = 15. The mantissa provides up to an additional 6.02dB of gain. The gain in dB from the mantissa is: 20log 10 [1+(X)2 -14], where X is the fractional part of the mantissa interpreted as an unsigned integer ranging from 0 to 214 - 1. Thus, the AGC multiplier / shifter transfer function is expressed as: AGC Mult/Shift Gain = 2N [1+ (X)2-14] where N, the shifter exponent, has a range of 022
HSP50216
AGC LOOP FILTER
P
AGC ERROR SCALING (RANGE = -2.18344 TO 2.18344)
AGC ERROR DETECTOR
REGISTER
LIMITER
SERIAL OUT
19
MSB = 0
M U X
+
28
SHIFT
16 MANTISSA 4
REGISTER
16 P (11 MANTISSA 4 EXPONENT)
EXP MSB = 0 LIMIT DET EN AGC LOAD AGCGNSEL
4
AGC REGISTER 0
AGC REGISTER 1
EXP
18
4
EXP=2NNNN
16 MANTISSA = 01.XXXXXXXXXXXXXX LIMIT DET MAGNITUDE (RANGE = 0 TO 2.32887) 16 (RANGE = 0 TO 1) LIMITER
SHIFTER
24 IFIR
24
24 IAGC LIMIT DET CARTESIAN TO POLAR COORDINATE CONVERTER (G = 1.64676)
SHIFTER
QFIR
LIMITER
24
24
24 QAGC
AGC MULTIPLIER/SHIFTER
Controlled via microprocessor interface.
FIGURE 1. AGC FUNCTIONAL BLOCK DIAGRAM
In dB, this can be expressed as: (AGC Mult/Shift Gain)dB = 20 log10(2N[1 + (X)2-14]) The full AGC range of the multiplier / shifter is from 0 dB to 20log 10 [1+(2 14 -1)2 -14 ] + 20log 10 [2 15 ] = 96.329 dB. The 16 bit resolution of the mantissa provides a theoretical AM modulation level of -96dBc (depending on loop gain, settling mode and SNR). This effectively eliminates AM spurious caused by the AGC resolution. The Cartesian to polar coordinate converter accepts I and Q data and generates magnitude and phase data. The magnitude output is determined by the equation:
r = 1.64676 I + Q
2 2
where the magnitude limits are determined by the maximum I and Q signal levels into the Cartesian to polar converter. Taking fractional 2's complement representation, magnitude ranges from 0 to 2.329, where the maximum output is
r = 1.64676 1 + 1
2 2
= 1.64676x1.414 = 2.329
The AGC loop feedback path consists of an error detector, error scaling, and an AGC loop filter. The error detector subtracts the magnitude output of the coordinate converter from the programmable AGC THRESHOLD value. The AGC THRESHOLD value is set in IWA register *012h and is equal to 1.64676 times the desired magnitude of the I1/Q1 output. Note that the MSB is always zero. The range of the AGC THRESHOLD value is 0 to +3.9999. The AGC Error Detector output has the identical range.
23
EXP MAN LOOP GAIN 1
LOOP GAIN 0
UPPER LIMIT LOWER LIMIT
MAN
UNSIGNED THRESHOLD STT.TTTTTTTTTTTTT (S = 0)
HSP50216
The loop gain register values adjust the response / settling time of the AGC loop. The loop gain is set in the AGC Error Scaling circuitry, using four values in two sets of programmable mantissa and exponent pairs (see IWA register *010h). Each set has both an attack and a decay gain. This allows asymmetric adjustment for applications such as VOX systems where the signal turns on and off. In these applications, the gains would be set for fast attack and slow decay so that the part decreases the gain quickly when the signal turns on, but increases the gain slowly when the signal turns off (in anticipation of it turning back on shortly). For fixed gains, either set the upper and lower AGC limits to the same value, or set the limits to minimum and maximum gains and set the AGC attack and decay loop gains to zero. The mantissa, M, is a 4-bit value which weights the loop filter input from 0.0 to 15 / 24 = 0.9375. The exponent, E, defines a shift factor that provides additional weighting from 2 0 to 2-15. Together the mantissa and exponent define the loop gain as given by, AGC Loop Gain = MLG 2-4 2-(15-ELG) where M LG is a 4-bit binary mantissa value ranging from 0 to 15, and E LG is a 4-bit binary exponent value ranging from 0 to 15. The composite (shifter and multiplier) AGC scaling Gain range is from 0.0000 to 2.329(0.9375)2 0 = 0.0000 to 2.18344. The scaled gain error can range (depending on threshold) from 0 to 2.18344, which maps to a "gain change per sample" range of 0 to 3.275dB / sample. The AGC attack and decay gain mantissa and exponent values for loop gains 0 and 1 are programmed into IWA register *010h. The PDC provides for the storing of two values of AGC attack and decay scaling gains to allow for quick adjustment of the loop gain by simply setting IWA register *013h bits 9 and 10 accordingly. Possible applications include acquisition / tracking, no burst present / burst present, strong signal / weak signal, track / hold, or fast / slow AGC values. The AGC loop filter consists of an accumulator with a built in limiting function. The maximum and minimum AGC gain limits are provided to keep the gain within a specified range and are programmed by 16-bit upper and lower limits using the following the equation: AGC Gain Limit = (1 + mAGC 2-12) 2e (AGC Gain Limit)dB = (6.02)(eeee) + 20 log(1.0+0.mmmm mmmm mmmm) where m is a 12-bit mantissa value between 0 and 4095, and e is the 4-bit exponent ranging from 0 to 15. IWA register *011h Bits 31:16 are used for programming the upper limit, while bits 15:0 are used to program the lower limit. The format for these limit values are: (31:16) or (15:0): E E E E M M M M M M M M M M M M for a gain of 0 1. M M M M M M M M M M M M * 2 E E E E and the possible range of AGC limits from the previous equations is 0 to 96.328dB. The bit weightings for the AGC Loop Feedback elements are detailed in Table 51. Using AGC loop gain, the AGC range, and expected error detector output, the gain adjustments per output sample for the loop filter section of the digital AGC can be given by AGC Slew Rate = (1.5 dB) (THRESHOLD - (MAG * 1.64676)) x (MLG) (2-4) (2-(15 - ELG)) The loop gain determines the growth rate of the sum in the loop accumulator which, in turn, determines how quickly the AGC gain scales the output to the threshold value. Since the log of the gain response is roughly linear, the loop response can be approximated by multiplying the maximum AGC gain error by the loop gain. The expected range for the AGC rate is ~ 0.000106 to 3.275dB / output sample time for a threshold of 1/2 scale. For a full scale error, the minimum non-zero AGC slew rate would be approximately 0.0002dB / output or 20dB / sec at 100ksps. The maximum gain would be 6dB / output. This much gain, however, would probably result in significant AM on the output. The maximum AGC Response is given by: AGC ResponseMax = (Input)(Cart/Polar Gain)(Error Det. Gain)(AGC Loop Gain)(AGC Output Weighting) Since the AGC error is scaled to adjust the gain, the loop settles asymptotically to its final value. The loop settles to the mean of the signal. For example, if MLG = 0101 and ELG = 1100, the AGC Loop Gain = 0.3125 * 2 -7. The loop gain mantissas and exponents are set in IWA register *010h, with IWA register *013h selecting loop gain 0 or 1 and the settling mode. In the HSP50216, a SYNCI signal will clear the AGC loop filter accumulator if GWA register F802h bit 27 is set. The settling mode of the AGC forces either the mean or the median of the signal magnitude error to zero, as selected by IWA register *013h bit 8. For mean mode, the gain error is scaled and used to adjust the gain up or down. This proportional scaling mode causes the AGC to settle to the final gain value asymptotically. This AGC settling mode is preferred in many applications because the loop gain adjustments get smaller and smaller as the loop settles, reducing any AM distortion caused by the AGC. With this AGC settling mode, the proportional gain error causes the loop to settle more slowly if the threshold is small. This is because the maximum value of the threshold minus the magnitude is smaller. Also, the settling can be asymmetric, where the loop may settle faster for "over range" signals than for "under range" signals (or vice versa). In some applications, such as burst signals or TDMA signals, a very fast settling time and/or a more predictable settling time is desired. The AGC may be turned off or slowed down after an initial AGC settling period.
24
HSP50216
The median mode minimizes the settling time. This mode uses a fixed gain adjustment with only the direction of the adjustment controlled by the gain error. This makes the settling time independent of the signal level. For example, if the loop is set to adjust 0.5dB per output sample, the loop gain can slew up or down by 16dB in 16 symbol times, assuming a 2 samples per symbol output sample rate. This is called a median settling mode because the loop settles to where there is an equal number of magnitude samples above and below the threshold. The disadvantage of this mode is that the loop will have a wander (dither) equal to the programmed step size. For this reason, it is advisable to set one loop gain for fast settling at the beginning of the burst and the second loop gain for small adjustments during tracking. In the median mode, the maximum gain step is approximately 3dB / output. The step is fixed (it does not decrease as the error decreases) so a large gain will cause AM on the output at least that large. The gain should be lowered after the settling. The fixed gain step is set by the programmable AGC loop gain register IWA *010h. For median mode, The AGC gain limits register sets the minimum and maximum limits on the AGC gain. The total AGC gain range is 96dB, but only a portion of the range should be needed for most applications. For example, with a 16-bit output to a processor, the 16 bits may be sufficient for all but 24dB of the total input range possible. The AGC would only need to have a range of 24dB. This allows faster settling and the AGC would be at its maximum gain limit except when a high power signal was received. The AGC may be disabled by setting both limits to the same value. The median settling mode is enabled by setting IWA register *013h bit 8 to 0 while the mean loop settling mode is selected by setting bit 8 to 1.
TABLE 1. MAG/PHASE BIT WEIGHTING BIT 23 (MSB) 22 21 20 19 18 17 16 15 14 13 12 11 10 9 8 7 6 5 4 3 2 1 0 (LSB)
400000 3ff fff Q 7fffff
MAGNITUDE 22 21 20 2-1 2-2 2-3 2-4 2-5 2-6 2-7 2-8 2-9 2-10 2-11 2-12 2-13 2-14 2-15 2-16 2-17 2-18 2-19 2-20 2-21 180 90 45 22.5
PHASE (o)
11.25 5.625 2.8125 1.40625 0.703125 0.3515625 0.17578125 0.087890625 0.043945312 0.021972656 0.010986328 0.005483164 0.002741582 0.001370791 0.0006853955 0.00034269775 0.00017134887 0.00008567444 0.00004283722 0.00002141861
/2 400000 3fffff Q 7fffff 800000 I 000000 0 ffffff
+/2
Cartesian to Polar Converter
The Cartesian to Polar converter computes the magnitude and phase of the I/Q vector. The I and Q inputs are 24 bits. The converter phase output is 24 bits, MSB's routed to the output formatter and all 24 bits routed to the frequency discriminator. The 24-bit output phase can be interpreted either as two's complement (-0.5 to approximately 0.5) or unsigned (0.0 to approximately 1.0), as shown in Figure 2. The phase conversion gain is 1/2. The phase resolution is 24 bits. The 24-bit magnitude is unsigned binary format with a range from 0 to 2.32. The magnitude conversion gain is 1.64676. The magnitude resolution is 24 bits. The MSB is always zero. Table 1 details the phase and magnitude weighting for the 16 bits output from the PDC.
800000
I 000000 0 ffffff bfffff c00000
-/2
bfffff c00000 3/2
FIGURE 2. PHASE BIT MAPPING OF COORDINATE CONVERTER OUTPUT
The magnitude and phase computation requires 17 clocks for full precision. At the end of the 17 clocks, the magnitude and phase are latched into a register to be held for the next stage, either the output formatter or frequency discriminator. If a new input sample arrives before the end of the 17 cycles, the results of the computations up until that time, are latched. This latching means that an increase in speed causes only a decrease in resolution. Table 2 details the exact resolution that can be obtained with a fixed number of clock cycles up to the required 17. The input magnitude and phase errors induced by normal SNR values will almost always be worse than the Cartesian to Polar conversion.
25
HSP50216
TABLE 2. MAG/PHASE ACCURACY vs CLOCK CYCLES MAGNITUDE ERROR (% fS) 0.065 0.016 0.004 <0.004 <0.004 <0.004 <0.004 <0.004 <0.004 <0.004 <0.004 <0.004 PHASE ERROR (DEG.) 3.5 1.8 0.9 0.45 0.22 0.11 0.056 0.028 0.014 0.007 0.0035 0.00175 PHASE ERROR (% fS) 2 1 0.5 0.25 0.12 0.062 0.03 0.016 0.008 0.004 0.002 0.001
CLOCKS 6 7 8 9 10 11 12 13 14 15 16 17
Assumes 180o = fS.
The enable signal for gating data into the coordinate converter is either the AGC data ready signal or the resampler data ready signal. If the resampler is bypassed, the AGC data ready signal is used and there is a delay of 6 clock cycles between the FIR data being ready and the coordinate converter block sampling it. If the resampler is enabled, its data ready signal will be delayed by 6 clocks (for the AGC) plus the compute delay of the resampler block. This may cause the I/Q to |r|/ output sample alignment to shift with the decimation. For this reason, it is recommended that the resampler/halfband filter block be bypassed when using this new data path.
26
HSP50216 Serial Data Output Formatter Section
OUTPUT SECTION
ZERO I1 Q1 MAG PHASE I2 Q2 GAIN STROBE ZERO PARALLEL TO SERIAL DELAY & &O &R & SD2x R E G M U X & FIXED TO FLOAT ROUND & SEQUENCER 1 &O &R & M U X PARALLEL TO SERIAL &O &R & SD1x
SYNC GEN
SYNCx
M U X
ROUND
SEQUENCER 2
M U X
16
TO P INTERFACE
NOTE: Each serial output has 7 time slots. Each slot can contain I1, Q1, I2, Q2, Mag, phase or d/dt. AGC gain, or zeros. Each slot can be 4, 6, 8, 10, 12, 16, 20, 24, or 32 (24 + 8 zeros) bits or disabled. Output 1 can also be 32-bit floating point. Slots can be disabled. A disabled slot will be one clock wide if there are other active slots following. A sync can be asserted with any or all slots following. A sync can be asserted with any or all slots in output 1. The serial output can be delayed from 0 to 4095 serial clock periods from the input strobe. The serial outputs are always MSB first. The sync position applies to all time slots and can be one clock prior to the first data bit, aligned with the first data bit, or one clock after the last data bit.
Serial Data Output Control Register
The serial data output control register contains sync position and polarity (SYNCA, B, C or D), channel multiplexing, and scaling controls for the SD1x and SD2x (x = A, B, C or D) serial outputs (see Microprocessor Interface section, Table 23, "SERIAL DATA OUTPUT CONTROL REGISTER (IWA = *014h)," on page 37).
Channel Routing Mask
The multiplexing mask bits for each channel (see Microprocessor Interface section, Table 23, IWA *014h bits 19:16 for SD1x or bits 15:12 for SD2x) can be used to enable that channel's output to any of the four serial outputs. These bits control the AND gates that mask off the channels, so a zero disables the channel's connection to that output.
To configure more than one channel's output onto a serial data output, the SD1 serial outputs and syncs from each channel (0,1, 2 and 3) are brought to each of the SD1 serial output sections and the SD2 serial outputs are brought to each of the SD2 serial output sections (the syncs are only associated with the SD1 serial outputs). There, the four outputs are AND-ed with the multiplexing mask programmed in the serial data output control registers of channels 0 through 3 and OR-ed together. By gating off the channels that are not wanted and delaying the data from each desired channel appropriately, the channels can be multiplexed into a common serial output stream. It should be noted that in order to multiplex multiple channels onto a single serial data stream the channels to be multiplexed must be synchronous.
27
HSP50216
Serial Data Output Time Slot Content/Format Registers
These four registers are used to program the content and format of the serial data output sequence time slots (see Microprocessor Interface section: Table 24, "SERIAL DATA OUTPUT 1 CONTENT/FORMAT REGISTER 1 (IWA = *015h)," on page 39 through Table 27, "SERIAL DATA OUTPUT 2 CONTENT/FORMAT REGISTER 2 (IWA = *018h)," on page 40). There are seven data time slots that make up a serial data output stream. The number of data bits and data format of each slot is programmable as well as whether there will be a sync generated with the time slot (the syncs are only associated with the SD1 serial outputs). Any of seven types of data or zeros can be chosen for each time slot. Eight bits are used to specify the content and format of each slot. As an example, suppose we wanted to output 32-bit I and Q values from channels 0 and 1 into the SD1A serial data output stream, we would program the following settings in the channel's serial data output control and content/format registers: Channel 0: delay = 0 (IWA = 0014h, bits 11:0 = 0); first data time slot = I, 32-bit, sync pulse generated (IWA = 0015h, bits 7:0 = 0xC9); second data time slot = Q, 32-bit, no sync pulse (IWA = 0015h, bits 15:8 = 0x4A); third through seventh data time slot = zero and no sync, (IWA = 0015h, bits 31:16 = 0 and IWA = 0016h, bits 31:0 = 0); enable the SD1A serial output for this channel in the serial routing mask (IWA = 0014h, bit 16 = 1). Channel 1: delay = 64 (IWA = 1014h, bits 11:0 = 0x40); first data time slot = I, 32-bit, sync pulse generated (IWA = 1015h, bits 7:0 = 0xC9); second data time slot = Q, 32-bit, no sync pulse (IWA = 1015h, bits 15:8 = 0x4A); third through seventh data time slot = zero and no sync, (IWA = 1015h, bits 31:16 = 0 and IWA = 1016h, bits 31:0 = 0); enable the SD1A serial output for this channel in the serial routing mask (IWA = 1014h, bit 16 = 1). The resulting order is CH0 I first, then CH0 Q, CH1 I, and CH1 Q with sync pulses generated in the I data slots. The position of the sync pulses relative to the data slot may be programmed with IWA register *014h bits 25:24. Setting delay = 64 offsets channel 1's 32 bit I and Q data by 64 clocks so that it immediately follows the 64 bits of data from channel 0. In this way channel 1's first and second time slots follow channel 0's second time slot. Instead of using the delay to offset channel 1's data, channel 0 could have been configured to output 32 bits of I in the fist slot, 32 bits of Q in the second slot, 32 bits of zeros in the third slot and 32 bits of zeros in the fourth slot. Channel 1 could then be configured to output 32 bits of zeros in the first and second slots, 32 bits of I in the third slot and 32 bits of Q in the fourth slot. As the channel outputs are OR'd together, the zero slots do not interfere with data slots. The HSP50216 Microprocessor (P) interface consists of a 16-bit bidirectional data bus, P(15:0), three address pins, ADD(2:0), a write strobe (WR), a read strobe (RD) and a chip enable (CE). Indirect addressing is used for control and configuration of the HSP50216. The control and configuration data to be loaded is first written to a 32-bit holding register at direct (external) addresses ADD(2:0) = 0 and 1, 16 bits at a time. The data is then transferred to the target register, synchronous to the clock, by writing the indirect (internal) address of the target register to direct (external) address 2, ADD(2:0) = 2. The interface generates a synchronous one clock cycle wide strobe to transfer the data contained in the holding register to the target register. The synchronization and write process requires 4 clock periods. New data should not be written to the holding register until after the synchronization period is over.
28
HSP50216 Microprocessor Interface
15:0 MUX 3210 31:16 31:0 INTERNAL READ DATA BUS M U X E S
FROM OUTPUT FIFO STATUS
RD L A T C H
P(15:0)
en >
R E G
15:0
31:0 INTERNAL WRITE DATA BUS
WR
en >
R E G R E G
31:16
INTERNAL ADDRESS BUS
A(2:0)
D E C O D E
=0 =1 = 2 or 3 =2
en > RST
AND > F F > F F > F F > F F
G A T I N G
SYNC'd WR
CLK SPECIAL LOW METASTABILITY CELL CE (GATING NOT SHOWN)
TO TARGET REGISTERS
INTERNAL READ SIGNAL
Data reads can be direct, indirect or FIFO-like depending on the data that is being read. The status register is read directly at direct (external) address 3, ADD(2:0) = 3. Readback of internal registers and memories is indirect. The 16-bit indirect (internal) address of the desired read source is first written to direct (external) address 3, ADD(2:0) = 3, to select the data. The data can then be read at direct (external) addresses ADD(2:0) = 0 and 1 (bits 15:0 at address 0 and 31:16 at address 1). The data types available via the indirect read are listed in the Tables of Indirect Read Address (IRA) Registers. (Note that the PHold bit contained in the target register at Indirect Write Address (IWA) = *00Ah must be set to suspend the filter compute engine before the coefficient RAM and instruction bit fields can be written to or read from.) The HSP50216 output data from the four channels is available through the microprocessor interface as well as from the serial data outputs. A FIFO-like interface is used to read the output data through the microprocessor interface. When new output data is available, it is loaded into a FIFO in a user programmed order (for details on the programming order, see Tables of Global Write Address (GWA) Registers (GWA) = F820h - F83Fh). It can then be read, 16 bits at a time, at direct address 2, ADD(2:0) = 2. At the end
of each read, the FIFO counter is advanced to the next location. This allows a DMA controller to read all of the data with successive reads to a single direct address. No writes or other interaction is required. The FIFO counter is reset and reloaded by each interrupt signal, see GWA F802h. New data in the FIFO is also indicated in the status register located at direct address ADD(2:0) = 3 if a polled mode is preferred. The eight data types available, for each of the four channels, via this interface are: I(23:8), I(7:0)+8 Zeroes, Q(23:8), Q(7:0)+8 Zeroes, Mag(23:8), Mag(7:0)+8 Zeroes, Phase (15:0), and AGC (15:0). The upper bits of I, i.e., I(23:8), and Q, i.e., Q(23:8), are not rounded to 16 bits. This interface can read the data from all the channels that are synchronized. However, because a common FIFO is used and the FIFO is reset and reloaded by each interrupt, it cannot be used for asynchronous channels.
29
HSP50216
The direct address map for the microprocessor interface is shown in the TABLE OF MICROPROCESSOR DIRECT READ/WRITE ADDRESSES and the procedures for reading and writing to this interface are provided below. The bit field details for each indirect read and write address are provided in the Table of Indirect Read Address (IRA) Registers, Tables of Indirect Write Address (IWA) Registers (Tables 3 34) and Tables of Global Write Address (GWA) Registers (GWA) Registers (Tables 35 - 45). 3. Data can then be read, 16 bits at a time, at direct address 2, ADD(2:0) = 2. 4. Repeat step 3 for desired number of words. 5. Go to step 2.
To Read Instruction/Coefficient Values:
1. Put the filter compute engine of the desired channel into the hold mode by setting bit 31 of the Filter Compute Engine / Resampler Control register located at IWA = *00Ah (Note: The * is equal to 0, 1, 2 or 3 depending on the channel being addressed). 2. Write the Indirect Read Address (IRA) of the internal RAM/ROM location being addressed to direct address ADD(2:0) = 3. 3. Wait 4 clock cycles. 4. Read the data at direct address ADD(2:0) = 0 and 1. 5. After all the data has been read, set the PHold bit back low.
P Read/Write Procedures
To Write to the Internal Registers:
1. Load the indirect write holding registers at direct address ADD(2:0) = 0 and 1 with the data for the internal register (16 or 32 bits depending on the internal register being addressed). 2. Write the Indirect Write Address of the internal register being addressed to direct address ADD(2:0) = 2 (Note: A write strobe to transfer the contents of the Indirect Write Holding Register into the Target Register specified by the Indirect Address will be generated internally). 3. Wait 4 clock cycles before performing the next write to the indirect write holding registers.
Recommended HSP50216 configuration procedure following a hardware reset (i.e. RESETb is pulsed low):
1. Load Global Write Address registers GWA F800 - GWA F808 and GWA F820 - GWA F83F. 2. For each signal processing channel (0-3): a. Set mPHold bit located at Indirect Write Address register IWA *00A - 31. b. Load Filter Compute Engine Instruction RAMS. c. Load Filter Compute Engine Coefficient RAMS. d. Load IWA registers *000 - *019. (Clear the mPHold bit in register IWA *00A - 31). e. Wait 32 clocks (CLK) for the reset to complete in the Filter Compute Engine. 3. Generate a SYNCI to enable the input data or to synchronize the processing to external events or generate a SYNCO by writing to GWA F809.
NOTE: For the latter method, the SYNCO pin must be connected to the SYNCI pin.
To Write to the Internal Instruction/Coefficient RAMs:
1. Put the filter compute engine of the desired channel into the hold mode by setting bit 31 of the Filter Compute Engine / Resampler Control register located at IWA = *00Ah (Note: The * is equal to 0, 1, 2 or 3 depending on the channel being addressed). By setting bit 31 all FIR processing for the channel addressed will be stopped. 2. Load the indirect write holding registers at direct address ADD(2:0) = 0 and 1 with the data for the internal RAM location. 3. Write the Indirect Write Address of the internal RAM location being addressed to direct address ADD(2:0) = 2 (Note: A write strobe to transfer the contents of the Indirect Write Holding Register into the RAM location specified by the Indirect Address will be generated internally). 4. Wait 4 clock cycles before performing the next write to the indirect write holding registers. 5. After all data has been loaded, set the PHold bit back low.
Recommended HSP50216 Channel Reconfiguration Procedure:
1. Disable the serial output for the desired channel in register GWA F801 - 3, 2, 1 or 0. 2. Disable the interrupts from the channel in register GWA F802 - 31, 23, 15, or 7. 3. Set the mPHold bit in register IWA *00A - 31 to give the processor access to the Filter Compute Engine Instruction RAMS and Coefficient RAMS. 4. Load the new filter configuration. 5. Load any other channel registers.
To Read Internal Registers:
1. Write the Indirect Read Address of the internal register being addressed to direct address ADD(2:0) = 3. 2. Perform a read of the Indirect Read Holding Registers at direct address ADD(2:0) = 0 and 1.
To Read Data Outputs:
1. Set up the P FIFO Read Order Control Register (located at Global Write Address (GWA) = F820h - F83Fh). 2. Wait for interrupt or check flag.
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6. Clear the mPHold bit in register IWA *00A - 31. 7. Do a software channel reset by writing to IWA *019. 8. Enable the serial outputs (GWA F801) and interrupts (GWA F802). 9. Generate a SYNCI to enable the input data or to synchronize the processing to external events or generate a SYNCO by writing to GWA F809.
NOTE: For the latter method, the SYNCO pin must be connected to the SYNCI pin.
TABLE OF MICROPROCESSOR DIRECT READ/WRITE ADDRESSES ADD(2:0) 0 1 2 PINS WR WR WR REGISTER DESCRIPTION Indirect Write Holding Register, Bits 15:0. Indirect Write Holding Register, Bits 31:16. Indirect Write Address Register for Internal Target Register (Generates a write strobe to transfer contents of the Write Holding Register into the Target Register specified by the Indirect Address, see also Table of Indirect Read Address (IRA) Registers). Indirect Read Address Register (Used to select the Read source of data - uses the same register as Direct Address 2 but generates a read strobe (for RAMs and AGC) as needed instead of a write strobe). Indirect Read, Bits 15:0. Indirect Read, Bits 31:0f. Read Register (FIFO) - Reads FIFO data from output section (This location reads output data in the order loaded in Global Control Indirect Address Registers F820-F83F. The FIFO is automatically incremented to the next data location at the end of each read). Status Register P(15:0) 15:12 11:6 Unused. Read non-bus input pins (ENIx, RESET, SYNCI). 11 RESET (Note: This bit is inverted with respect to the RESET input pin). 10 ENIA. 9 ENIB. 8 ENIC. 7 ENID. 6 SYNCI. 5:2 1 0 Mask revision number. Level detector integration done. Active high. New FIFO output data available (used for polling mode vs interrupt mode) Active low. BIT DESCRIPTION
3 0 1 2
WR RD RD RD
3
RD
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HSP50216
Tables of Indirect Write Address (IWA) Registers
NOTE: These Indirect Write Addresses are repeated for each channel. In the addresses below, the * field is the channel select nibble. These bits of the Indirect Address select the target channel register for the data. Values of 0 through 3 and F are valid. A channel select nibble value of F is a special case which writes the data to the same location in each of the four channels simultaneously. TABLE 3. CHANNEL INPUT SELECT/FORMAT REGISTER (IWA = *000h) P(15:0) 15:13 FUNCTION Channel Input Source Selection - Selects as the data input for the channel specified in the Indirect Address either A(15:0), B(15:0), C(15:0), D(15:0) or the P Test Input register as shown below: 15:13 000 001 010 011 100 12 Source Selected A(15:0) B(15:0) C(15:0) D(15:0) P Test input register. This is provided for testing and to zero the input data bus when a channel is not in use. The Global Write Address register for the P Test input register is F807h.
P Test Register input enable selection: 1 0 Bit 11 of this register is used as the input enable. A one clock wide pulse generated on each write to lGWA F808h is used as the input enable.
Select 0 to write test data into the part. Select 1 to input a constant or to disable the input for minimum power dissipation when an NCO/mixer/CIC section is unused. 11 P input enable. When bit 12 is set, this bit is the input enable for the P Test Register input. Active low: 0 1 10 Enabled Disabled.
Parallel Data Input Format: 0 1 Two's complement (-full scale = 1000...0000, zero = 0000...0000, +full scale = 0111...1111). Offset binary (-full scale = 0000...0000, zero = 1000...0000, +full scale = 1111...1111).
9
Fixed/Floating point: 0 1 Fixed point. Floating point. The 16-bit input bus is divided into mantissa and exponent bits grouped either 13/3 or 14/2 depending on bits 8 and 7. See text.
8:7
Floating point mantissa size select. The 16-bit data input is grouped as a 13/3 or 14/2 mantissa/exponent word. These control bits select the mantissa/exponent grouping, add an offset to the exponent and set the shift control saturation level: 00 01 10 11 11/3: bits 15:5 are mantissa, 2:0 are exponent. 12/3: bits 15:4 are mantissa, 2:0 are exponent. 13/3: bits 15:3 are mantissa, 2:0 are exponent. 14/2: bits 15:2 are mantissa, 1:0 are exponent.
See the exponent tables contained in the Input Select/Format Block section. 6:4 De-multiplex control. These control bits are provided to select a channel from a group of multiplexed channels. Up to 8 multiplexed data streams can be demultiplexed. These control bits select how many clocks after the ENIx signal to wait before taking the input sample. ENIx should be asserted for one clock period and aligned with the first channel of the multiplexed data set. For example, if four streams are multiplexed at half the clock rate, ENIx would align with the first clock period of the first stream, the second would start two clocks later, the next 4 clocks after ENIx, etc. The samples are aligned with ENIx (zero delay) at the input of the NCO/Mixer/CIC stage at the next ENIx. 000 111 Zero delay 7 clock periods of delay.
All values from 0 through 7 are valid. 3 Interpolated/Gated Mode Select: 0 1 Gated. The carrier NCO and CIC are updated once per clock when ENIx is asserted. Interpolated. The CIC is updated every clock. The carrier NCO is updated once per clock when ENIx is asserted. The input is zeroed when ENIx is high.
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TABLE 3. CHANNEL INPUT SELECT/FORMAT REGISTER (IWA = *000h) (Continued) P(15:0) 2 1 0 FUNCTION Enable COF/COFSYNC inputs. When set, this bit enables two bits from the D(15:0) input data bus to be used as a carrier offset frequency input. Enable SOF/SOFSYNC inputs. When set, this bit enables two bits from the D(15:0) input data bus to be used as a resampler offset frequency input. Enable PN. When set, A PN code, weighted by the gain in location *001, is added to the input samples at the output of the mixer. TABLE 4. PN GAIN REGISTER (IWA = *001h) P(31:0) 31:16 15:0 Reserved, set to all 0's. PN generator gain register. This input is provided to reduce the sensitivity of the receiver. A PN code, weighted by the value in this location, is added to the data at the output of the mixer. Adding noise has the effect of increasing the receiver noise figure. One reason to do this would be to decrease the basestation cell size in small steps. This method is very accurate and repeatable and can be done on a FDM channel by channel basis. It does, however, reduce the overall dynamic range. An alternate way is to add attenuation at the RF and adjust the whole range upward. This does not reduce the overall range but only shift it, with the shift being done on all channels simultaneously. TABLE 5. CIC DECIMATION FACTOR REGISTER (IWA = *002h) P(15:0) 15:0 Load with the desired CIC decimation factor minus 1. TABLE 6. CIC DESTINATION FIR AND OUTPUT ENABLE/DISABLE REGISTER (IWA = *003h) P(15:0) 15:6 5:1 0 FUNCTION Set to zero. CIC output destination (FIR # in FIR processor). Usually set to 00001. CIC output enable. Active high. When low, the data writes from the CIC to the filter compute engine are inhibited. TABLE 7. CARRIER NCO/CIC CONTROL REGISTER (IWA = *004h) P(31:0) 31:19 18:14 Reserved, set to zero. CIC barrel shift control. 00000 is the minimum shift factor and 11111 is maximum shift factor. This compensates for the CIC filter gain of RN, where N is the number of enabled CIC stages and R is the CIC decimation factor. The equation used to compute the shift factor is: Shift Factor = 45 - Ceiling(log2(RN)). Examples: N R 5 5 13:9 512 8 Shift Factor 0 30 FUNCTION FUNCTION FUNCTION
CIC stage bypasses. The integrator/comb pairs are numbered 1 through 5 with 1 being the first integrator and first comb. Bit 13 bypasses the first integrator/comb pair, bit 12 bypasses the second, etc. The first integrator is the largest. Typically, the stages are enabled starting with stage 1 for maximum decimation range. Carrier phase shift. Phase shifts of N*(/4), N = 0 to 7. Clear feedback (test signal or for mixer bypass). NCO clear feedback on load. Update frequency on SYNCI. Redundant. Set to1. See Table 37, "RESET/SYNC/INTERRUPT SOURCE SELECTION REGISTER (GWA = F802h)," on page 43.
8:6 5 4 3
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TABLE 7. CARRIER NCO/CIC CONTROL REGISTER (IWA = *004h) (Continued) P(31:0) 2:1 FUNCTION Number of Carrier Offset Frequency (COF) serial input bits. The format is 2's complement, early SYNC, MSB first: 00 01 10 11 0 8 16 24 32
Enable serial carrier offset frequency (zeros the data already loaded via the COF/COFSYNC pins). To disable the COF shifting see IWA register *000h. TABLE 8. CARRIER NCO CENTER FREQUENCY REGISTER (IWA = *005h)
P(31:0) 31:0 Carrier Center Frequency (CCF):
FUNCTION
This is the frequency control for the carrier NCO. The center frequency control is double buffered. The contents of this register are transferred to the active register on a write to the CCFStrobe location or on a SYNCI (if load on SYNCI is enabled). The carrier center frequency is: CCF*fCLK/(232). CCF is a twos complement number and has a range of -231 to (231-1). fCLK is the input sample rate (ENIx assertion rate) for gated mode and the clock rate for interpolated mode. TABLE 9. CARRIER NCO CENTER FREQUENCY UPDATE STROBE REGISTER (IWA = *006h) P(15:0) N/A FUNCTION Writing to this address generates a strobe that transfers the CCF value to the active frequency register. The transfer to the active register can also be done using the SYNCI pin to synchronize the transfer in multiple parts or to synchronize to an external event. The value in the active register can be read at this address (the center frequency control before the serially loaded offset value is added). To read the value, either write this address to A(1:0) = 11 and then read at A(1:0) = 00 and 01, or read the value at A(1:0) = 00 and 01 after writing to this address and before writing a new address to either A(1:0) = 10 or 11. TABLE 10. TIMING NCO FREQUENCY CONTROL REGISTER, MSW (IWA = *007h) P(31:0) 31:0 FUNCTION These are the upper 32 bits of the 56-bit timing (resampler) NCO center frequency control. TABLE 11. TIMING NCO FREQUENCY CONTROL REGISTER, LSW (IWA = *008h) P(31:0) 31:8 7:0 FUNCTION These are the lower 24 bits of the 56-bit timing (resampler) NCO center frequency control. Unused, set to zero. TABLE 12. TIMING NCO CENTER FREQUENCY LOAD STROBE REGISTER (IWA = *009h) P(31:0) FUNCTION
N/A for WR A write to this location will update the resampler NCO center frequency. The upper 32 bits of the active register can be read at this 31:0 for RD address. TABLE 13. FILTER COMPUTE ENGINE/RESAMPLER CONTROL REGISTER (IWA = *00Ah) P(31:0) 31 FUNCTION PHold. When set, this bit stops the filter compute engine and allows the P access to the instruction and coefficient RAMs for reading and writing. On the high to low transition, the filter compute engine is reset (the read and write pointers are reset and the instruction at location 31 is fetched). PShiftZeroB. This bit, when set to zero, disables the coefficient shift bits (bits 9:8 of the master register when coefficient loading). PEN Limit. This bit disables the data path saturation logic. Provided for test. Active high. Set to 0 to disable the normal ROM controlled limiting (ANDed with normal signal).
30 29
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TABLE 13. FILTER COMPUTE ENGINE/RESAMPLER CONTROL REGISTER (IWA = *00Ah) (Continued) P(31:0) 28:24 FUNCTION PZ(4:0). These bits, when set to zero, zero the corresponding read pointer address bits. This allows the pointers to be aliased, i.e., multiple filters can access and/or modify the same pointer. They are provided to change filters, coefficients or decimation over a sequence. Unused, set to 0. Timing (resampler) NCO ENsync. If this bit is set, the center frequency is updated on a SYNCI. Set to 1. RSRVRS(1:0). Set to 01. Beginning/End. This bit selects whether the resampler NCO is updated at the beginning of a FIR computation or at the end of each FIR output computation. Usually, the resampler will be updated once at the beginning of each resampler computation and this will be bit set to 1. 1 0 18 Once at the beginning of the FIR instruction. At the last tap of each of the instruction's FIR computations (once per output).
23 22 21:20 19
RSModeSelect. This bit selects whether the resampler is a phase shifter or a frequency shifter. 0 1 Phase shift. It uses the top 5-bits of the timing NCO frequency to determine a phase shift and disables feedback in the timing NCO phase accumulator -- effect of the resampler is a constant phase shift. Frequency shift. effect of the resampler is a change in the sample rate.
17 16
RSCO. This bit is provided to force the resampler NCO carry when using the resampler as a phase shifter rather than for a frequency shift. This bit must be set for phase shifting and cleared for frequency shifting. (The bit is Or-ed with the normal carry.) RS NCO clear phase accumulator feedback on load. When this bit is set, the feedback in the resampler NCO phase accumulator is zeroed whenever the center frequency word is updated. This forces the NCO to a known phase so the phase of multiple channels can be aligned. Force NCO load. This bit, when set, zeroes the feedback in the resampler NCO phase accumulator. This is provided for test or to use the resampler for phase instead of frequency shifting. Enable RS freq offset. This bit, when set, enables the serially loaded resampler offset frequency word. When zero, the offset is zeroed. To disable the shifting, see IWA register *000h. Serial input word size. These bits select the number of bits in the resampler offset frequency word (loaded serially via SOF/SOFSYNC). 00 01 10 11 8 bits 16 bits 24 bits 32 bits
15 14 13:12
11:0
FIFODelay. A FIFO is provided at the output of the filter compute engine to smooth the sample spacing when using the resampler or interpolation FIRs. In these filters, the outputs can be produced in bursts or with gaps. The FIFO takes the samples in and outputs them based on a counter timeout. If the FIFO is empty and the counter is at its terminal count (hold state), the data is passed through and the counter is reloaded. If the counter is not at terminal count, the data is held in the FIFO until the counter times out. The FIFO can hold up to 4 samples. The delay is programmed in clock periods. The value programmed is one less than the number of clocks of delay. Set to 0 for a delay of one (fall through). The delay should be programmed to slightly less than the desired spacing to prevent overflow. TABLE 14. FILTER START OFFSET REGISTER (IWA = *00Bh)
P(15:0) 13:9 8:0
FUNCTION RAM Instruction number to which the offset is applied. 0-31. Aliasing applies. Used for polyphase filters. Amount of offset. Offsets the data RAM address for filter #n. This is used to offset the channels from each other when breaking the processing up among multiple channels for polyphase filters. For example, four channels can receive the same data at 8 MSPS, filter and decimate by 8 to output at 1MHz. If the computations are offset by 2 samples each, then the outputs of the four channels can be multiplexed together to get an output sample rate of 4MSPS. With a 64MSPS clock, the composite filter could have more than 100 taps where a single channel would only be capable of around 24 taps at a 4MHz output. EXCEPT IN VERY RARE CIRCUMSTANCES, THIS VALUE SHOULD BE A NEGATIVE NUMBER.
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TABLE 15. WAIT THRESHOLD/DECREMENT VALUE REGISTER (IWA = *00Ch) P(31:0) 31 30 29:20 19:10 9:0 FUNCTION PTestBit. This bit is provided as a microprocessor controlled condition code for the filter compute engine for conditional execution or synchronous startup. Active high. Set to 0. Decrement value 1. Positive number. Decrement value 0. Positive number. Usually set equal to the Threshold (bits 9:0). Threshold. Number of samples needed to run a filter set and produce an output. TABLE 16. RESET WRITE POINTER OFFSET REGISTER (IWA = *00Dh) P(15:0) 15:9 8:0 FUNCTION Set to zero. This parameter is the offset between filter compute engine read and write pointers on filter compute engine reset. On reset, the read and write pointers for all the filters are loaded, the read pointer with zero and the write pointer with this value. Set to zero for a single filter and two for a multi-filter chain. TABLE 17. AGC GAIN LOAD REGISTER (IWA = *00Eh) P(15:0) 15:0 FUNCTION This location loads the AGC accumulator. If the loop attack/decay gain is set to zero and this value is within the AGC gain limits, the AGC will hold this value. If not, the AGC will be set to this gain (or to a limit) and then start to settle. format is 4 exponent bits (15:12), and 12 mantissa bits, (11:0). TABLE 18. AGC GAIN READ STROBE REGISTER (IWA = *00Fh) P(15:0) FUNCTION
15:0 Writing to this location will sample the AGC loop filter output (forward gain value) to stabilize it for reading. The value is read from for RD; this location after waiting the 4 clocks required for read synchronization. N/A for WR TABLE 19. AGC LOOP ATTACK/DECAY GAIN VALUES REGISTER (IWA = *010h) P(31:0) 31:24 23:16 15:8 7:0 FUNCTION Loop gain 0, decay gain value (signal decay, increase gain) 31:28 = EEEE (exponent), 27:24 = MMMM (mantissa). Loop gain 1, decay gain value 23:20 = EEEE (exponent), 19:16 = MMMM (mantissa). Loop gain 0, attack gain value (signal arrival, decrease gain) 15:12 = EEEE (exponent), 11:8 = MMMM (mantissa). Loop gain 1, attack gain value 7:4 = EEEE (exponent), 3:0 = MMMM (mantissa). TABLE 20. AGC GAIN LIMITS REGISTER (IWA = *011h) P(31:0) 31:16 15:0 Upper gain limit. See AGC section. Lower gain limit. See AGC section. TABLE 21. AGC THRESHOLD REGISTER (IWA = *012h) P(31:0) 16 15:0 FUNCTION Enables dphi/dt update for non-fed back data. Discriminator output is not filtered. AGC threshold. Equals 1.64676 times the desired magnitude of the I1/Q1 output. FUNCTION
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HSP50216
TABLE 22. AGC/DISCRIMINATOR CONTROL REGISTER (IWA = *013h) P(15:0) 15:11 10 9 Set to zero. P AGC loop gain select. Enable filter compute engine control of AGC loop gain. When this bit is set, bit 28 in the filter compute engine destination field selects which loop gain to use with that filter output's gain error. Setting bit 10 overrides this bit and forces a loop gain 1. 10:9 00 10 01 11 8 FUNCTION Loop Gain 0 (P controlled) Loop gain 1 (P controlled) Loop Gain controlled by filter compute engine Loop 1 (P override of filter compute engine) FUNCTION
Mean/Median. This bit controls the settling mode of the AGC. Mean mode settles to the mean of the signal and settles asymptotically to the final value. Median mode settles to the median and settles with a fixed step size. This mode settles faster and more predictably, but will have more AM after settling. 1 0 Mean mode Median mode
7 6 5
Set this bit to 1 to get a dphi/dt output without having to feedback through the filter compute engine. Unused. Set to zero. PhaseOutputSel 1 0 d/dt Phase
4:3 2:0
DiscShift(1:0). Shifts the phase up 0, 1, 2, or 3-bit positions, discarding the bits shifted off the top. This makes the phase modulo 360, 180, 90, or 45 degrees to remove PSK modulation. The resulting phase is 18 bits. DiscDelay(2:0). Sets the delay, in sample times, for the d/dt calculation. 000 111 1 8 TABLE 23. SERIAL DATA OUTPUT CONTROL REGISTER (IWA = *014h)
P(31:0) 31:29 28 Set to zero. Sync polarity 1 0 27:26 25:24
FUNCTION
Active low (low for one serial clock per word with a sync). Active high.
Reserved, set to zero. Sync position. This applies to all time slots in the serial output. The Sync programming is associated with the SD1x serial output data stream (x = A, B, C, or D). 00 01 1X Sync is asserted during the serial clock period prior to the first data bit of the serial word (early sync). Sync is asserted during the clock period following the last data bit of the word (late sync). Sync is asserted during the serial clock period of the first data bit of the serial word (coincident sync).
23:22
Reserved, set to zero.
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TABLE 23. SERIAL DATA OUTPUT CONTROL REGISTER (IWA = *014h) (Continued) P(31:0) 21:20 FUNCTION Magnitude output scale factor. The magnitude output of the cartesian to polar coordinate conversion has bits weighted as: 2(2 1 0.-1 -2 -3 -4 . . . ) The gain in the conversion is 0.82338. When using 16 bits, the range is such that the LSB has a weight of 0.00007 and the maximum output is 2.32, both after the conversion gain. This corresponds to an I/Q vector length of -83dBFS to +3dBFS. These control bits add gain (with saturation) for more resolution at the bottom of the scale. A code of 00 passes the magnitude unchanged, 01 shifts the magnitude up one bit position' 10 shifts by 2 positions and 11 shifts up three positions. The resulting bit weights and range (after conversion gain) for the unsigned numbers are: Code 00 01 10 11 Bit Weights 2 1 0 -1 -2 . . . -11 -12 -13 1 0 -1 -2 -3 . . . -12 -13 -14 0 -1 -2 -3 -4 . . . -13 -14 -15 -1 -2 -3 -4 -5 . . . -14 -15 -16 dBFS +3 to -83 +3 to -89 +1.7 to -95 -4.3 to -101
The upper limits on codes 00 and 01 are the same, but 01 has no leading zero. 19:16 Serial data output SD1 routing mask. 0 disables. 1 enables. Bit 16 17 18 19 15:12 Enabled Output Enables the serial output for this channel to pin SD1A. Enables the serial output for this channel to pin SD1B. Enables the serial output for this channel to pin SD1C. Enables the serial output for this channel to pin SD1D.
Serial data output SD2 routing mask. 0 disables. 1 enables. Bit 12 13 14 15 Enabled Output. Enables the serial output for this channel to pin SD2A. Enables the serial output for this channel to pin SD2B. Enables the serial output for this channel to pin SD2C. Enables the serial output for this channel to pin SD2D.
11:0
Output hold-off delay. This parameter adds additional delay from the output of the filter compute engine to start of the serial output stream for multiplexing channels. Load with the desired delay (0 = zero, 1 = one, 2 = two, etc.).
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TABLE 24. SERIAL DATA OUTPUT 1 CONTENT/FORMAT REGISTER 1 (IWA = *015h) P(31:0) 31:24 23:16 15:8 7:0 FUNCTION Fourth serial slot in Serial Data Output 1 (SD1x). x = A, B, C or D. See bits 7:0 for functional description of bits 31:24. Third serial slot in Serial Data Output 1 (SD1x). x = A, B, C or D. See bits 7:0 for functional description of bits 23:16. Second serial slot in Serial Data Output 1 (SD1x). x = A, B, C or D. See bits 7:0 for functional description of bits 15:8. First serial slot in Serial Data Output 1 (SD1x). x = A, B, C or D. Bit Function 7 6:3 0000 0001 0010 0011 0100 0101 0110 0111 1000 1001 1010 Sync generated. When set, a sync pulse is generated with the data slot (Serial Data Output 1 only, i.e., the sync is only associated with Output 1). Set to zero for Output 2, SD2x. Word width/format. All fixed point data is twos complement. The data is rounded (asymmetrically, with saturation) to the desired number of bits. 0-bit, fixed point (actually 1-bit position is used). 4-bit, fixed point. 6-bit, fixed point. 8-bit, fixed point. 10-bit, fixed point. 12-bit, fixed point. 16-bit, fixed point. 20-bit, fixed point. 24-bit, fixed point. 32-bit fixed (8 LSBs are zeroed). 32-bit, floating point, IEEE format. All other codes are invalid. Note: Floating point format is only available on the Serial Data Output 1. Code 1010 is invalid on Serial Data Output 2. 2:0 000 001 010 011 100 101 110 111 Data type Zeros I1 (data routed from FIFO and AGC path). Q1 (data routed from FIFO and AGC path). Magnitude of I1/Q1. Phase (or d/dt) of I1/Q1. I2 (data routed directly from the filter processor). Q2 (data routed directly from the filter processor). AGC gain of I1/Q1 path. The filter processor must be programmed appropriately to route the data to I1/Q1 or I2/Q2. NOTE: Disable a slot by setting the 8-bit word to 00h. When disabled, a slot still uses one clock period. If, for example, the slots are programmed to 16-bit, disabled, 16-bit, there would a one clock idle period between the two 16-bit data words. If a new data sample occurs before the current set of data has been output, the new data will preempt the output and the first slot of the new data will begin immediately. If a late sync was programmed, it will not occur. 0 1 2 3 4 5 6 7 8 9 ABCDEF 0 1 2 3 4 5 6 7 8 9 ABCDEF I, Q MAG PH AGC 0 1 2 3 4 5 6 7 8 9 0 1 2 3 4 5 6 7 8 9 0 1 2 3 ZZZZZZZZ Z 1 2 3 4 5 6 7 8 9 0 1 2 3 4 5 6 7 8 9 0 1 2 3 Z Z Z Z Z Z Z Z (MSB zero unless shifted) 0 1 2 3 4 5 6 7 8 9 0 1 2 3 4 5 6 7 ZZZZZZZZZZZZZZ Z 1 2 3 4 5 6 7 8 9 0 1 2 3 4 5 6 7 Z Z Z Z Z Z Z Z Z Z Z Z Z Z (MSB zeroed) TABLE 25. SERIAL DATA OUTPUT 1 CONTENT/FORMAT REGISTER 2 (IWA = *016h) P(31:0) 31:24 23:16 15:8 7:0 Set to zero. Seventh serial slot in Serial Data Output 1 (SD1x). x = A, B, C or D. See bits 7:0 of Table 24 for functional description of bits 23:16. Sixth serial slot in Serial Data Output 1 (SD1x). x = A, B, C or D. See bits 7:0 of Table 24 for functional description of bits 15:8. Fifth serial slot in Serial Data Output 1 (SD1x). x = A, B, C or D. See bits 7:0 of Table 24 for functional description of bits 7:0. FUNCTION
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HSP50216
TABLE 26. SERIAL DATA OUTPUT 2 CONTENT/FORMAT REGISTER 1 (IWA = *017h) P(31:0) 31:24 23:16 15:8 7:0 FUNCTION Fourth serial slot in Serial Data Output 2 (SD2x). x = A, B, C or D. See bits 7:0 of Table 24 for functional description of bits 23:16. Third serial slot in Serial Data Output 2 (SD2x). x = A, B, C or D. See bits 7:0 of Table 24 for functional description of bits 23:16. Second serial slot in Serial Data Output 2 (SD2x). x = A, B, C or D. See bits 7:0 of Table 24 for functional description of bits 15:8. First serial slot in Serial Data Output 2 (SD2x). x = A, B, C or D. See bits 7:0 of Table 24 for functional description of bits 7:0. TABLE 27. SERIAL DATA OUTPUT 2 CONTENT/FORMAT REGISTER 2 (IWA = *018h) P(31:0) 31:24 23:16 15:8 7:0 Set to zero Seventh serial slot in Serial Data Output 2 (SD2x). x = A, B, C or D. See bits 7:0 of Table 24 for functional description of bits 23:16. Sixth serial slot in Serial Data Output 2 (SD2x). x = A, B, C or D. See bits 7:0 of Table 24 for functional description of bits 15:8. Fifth serial slot in Serial Data Output 2 (SD2x). x = A, B, C or D. See bits 7:0 of Table 24 for functional description of bits 7:0. TABLE 28. SOFTWARE RESET REGISTER (IWA = *019h) P(15:0) N/A FUNCTION Writing to this location resets the following activities of the functional block indicated. Input Format/Select, NCO, Mixer and CIC. Clears any pending enable in each channel's input demultiplexer function, loads the CIC decimation counter (the load value is indeterminate if the decimation counter preload register has not been loaded), clears all processing enables (stops all processing in the data path, but does not clear the data path registers). Filter Compute Engine: Resets the Read/Write pointers, fetch instruction 31 and start the filter program execution. AGC: Resets the compute blocks in both the forward and loop filter blocks (any calculations in progress are lost). Cartesian-to-Polar Coordinate Converter: Resets the compute blocks (any calculations in progress are lost). FIFO: Resets counter (clears the FIFO, all data is lost). Resampler Timing NCO: Clears the slave (active) frequency registers and clears the phase accumulator. Output Section: Resets the serial output section (clears all registers, counters, and flags but does not clear the configuration registers). Self Test Control: Resets the self test control logic of the front end (Input Format/Select, NCO, Mixer, and CIC) and the back end (Filter Compute Engine, AGC, and Cartesian-to-Polar Coordinate Converter). TABLE 29. CHANNEL TIMING ADVANCE STROBE REGISTER (IWA = *01Ah) P(15:0) N/A FUNCTION Writing to this location inserts one extra data sample in the CIC to FIR path by repeating a sample. Used for shifting the FIR filter compute engine timing. TABLE 30. CHANNEL TIMING RETARD STROBE Register (IWA = *01Bh) P(15:0) N/A FUNCTION Writing to this location deletes one data sample in the CIC to FIR path. Used for shifting the FIR filter compute engine timing. FUNCTION
40
HSP50216
TABLE 31. FILTER COMPUTE ENGINE INSTRUCTION RAMS (IWA = *100h THROUGH *17Fh) P(31:0) 31:0 FUNCTION These locations in RAM are used to store the Filter Compute Engine instruction words. There are 128 bits per instruction word with each word consisting of condition code selects, FIR parameters and data routing controls. The filter compute engine is controlled by a simple sequencer supporting up to 32 steps where each step is defined by a 128 bit instruction word. The 128 bit instruction word is assigned to RAM memory in four 32 bit data writes through the Microprocessor Interface starting with the low 32 bits. Hence, 128 32-bit memory locations are required per channel to support the 32 steps of the Filter Sequencer. See the Filter Compute Engine and Filter Sequencer sections of the data sheet for more details. TABLE 32. FILTER COMPUTE ENGINE INSTRUCTION POINTER RAMS (IWA = *180h THROUGH *1FCh) P(15:0) FUNCTION
TABLE 33. FILTER COMPUTE ENGINE COEFFICIENT RAM1 (IWA = *440h THROUGH *47Fh) P(31:0) 31:0 FUNCTION These locations in RAM are used to store the 22-bit filter coefficients used by the Filter Compute Engine of each channel in implementing a FIR filter. The 22-bit FIR filter coefficients are loaded in the upper 22 bits of each 32-bit RAM location. The two LSBs of the second byte (bits 9:8 of the total 32 bits, 31:0) are the shift bits. These are set to zero if not used. The least significant byte (bits 7:0 of the total 32 bits, 31:0) are ignored. RAM1 address space allows for storage of 64 filter coefficients out of the total of 192 filter coefficient storage locations. See the Filter Compute Engine and Filter Sequencer sections of the data sheet for more details. TABLE 34. FILTER COMPUTE ENGINE COEFFICIENT RAM2 (IWA = *480h THROUGH *4FFh) P(31:0) 31:0 FUNCTION These locations in RAM are used to store the 22-bit filter coefficients used by the Filter Compute Engine of each channel in implementing a FIR filter. The 22-bit FIR filter coefficients are loaded in the upper 22 bits of each 32-bit RAM location. The two LSBs of the second byte (bits 9:8 of the total 32 bits, 31:0) are the shift bits. These are set to zero if not used. The least significant byte (bits 7:0 of the total 32 bits, 31:0) are ignored. RAM2 address space allows for storage of 128 filter coefficients out of the total of 192 filter coefficient storage locations. See the Filter Compute Engine and Filter Sequencer sections of the data sheet for more details.
41
HSP50216
Tables of Global Write Address (GWA) Registers
NOTE: These Global Write Addresses control global functions on the HSP50216, so they are not repeated for each channel. The top five address bits select this set of registers (F8XXh). TABLE 35. TEST CONTROL REGISTER (GWA = F800h) P(31:0) 31:17 FUNCTION These bits can be routed to the output pins by setting bit 16 below. The bit to pin mapping is: 31 = Intrpt 28 = SYNCA 24 = SD1A 20 = SD2A 30 = SYNCO 27 = SYNCB 23 = SD1B 19 = SD2B 29 = SERCLK (unless x1 CLK is selected) 26 = SYNCC 22 = SD1C 18 = SD2C 25 = SYNCD 21 = SD1D 17 = SD2D
This is provided for testing board level interconnects. To control the SERCLK output, a divided down clock must be selected in the serial clock control register (GWA = F803h). 16 15:10 9 8 7:4 3 2 This bit, when high, routes bits 31:17 to the output pins in place of the normal outputs. Bit 0 of this register must also be set to activate this function. Unused - set to zero. Set-up time to CLOCK adjust. Adjusting the delay trades set up time for hold time. This bit is used to best center the delay without a mask change. Set-up time to WRITE adjust. Adjusting the delay trades set up time for hold time. This bit is used to best center the delay without a mask change. These bits, when set, route the MSB of the SIN output of the channel's carrier NCO to the number 2 serial output pin in place of the normal output. 7=CH0 6=CH1 5=CH2 4=CH3. Offset I PN by XORing bit 10 of the PN generator with the output PN. Enable (223 - 1) PN generator. The PN signal that can be added to the mixer output of each channel is produced from a (223 - 1) sequence, a (215 - 1) sequence or both. Two separate generators are provided. The outputs of both are XORed together to extend the repeat period. Either or both generators can be disabled. The XORed output can further be XORed with a delayed version of the (223 - 1) sequence on the I channel to decorrelate it from the Q channel. Otherwise, the same sequence will be used on both I and Q. Enable (215 - 1) PN generator. Test mode. When asserted, this bit puts the chip into internal (self) test mode. TABLE 36. BUS ROUTING CONTROL REGISTER (GWA = F801h) P(31:0) 31:24 23:20 Unused - set to zero. Interrupt pulse width. The width of the interrupt pulse at the pin can be programmed to be from 1 to 15 clocks wide. Program with the desired number of clocks. (NOTE: The pulse counter is only reset with the RESET pin. If a channel is reset by software or a SYNCI, any interrupt pulse in process will finish). DataRdy delay (CH1 only). Test. From 1-8. CH1 or CH3 AGC to CH0 ext AGC. This bit selects whether the AGC loop filter output from CH1 or CH3 is routed to the external AGC gain input of CH0. 0=CH3, 1=CH1. CH3 ext source mux sel. These bits select whether the CH2 source mux, CIC2, or FIR2out is routed to the external input of FIR3. 0=CH2srcmux, 1=FIR2, 2=CIC2. CH2 ext source mux sel. This bit selects whether the CH1 external source mux or FIR1out is routed to the external input of FIR2. 0=CH1srcmux, 1=FIR1out. CH1 ext source mux sel. This bit selects whether the CIC0 output or FIR0out is routed to the external input of FIR1. 0=CIC0, 1=FIR0out. CH0 backend input sel. 0=CIC0, 1=CIC1 (test). CH1 backend input sel 0=CIC1, 1=CH1 ext src mux. CH2 backend input sel 0=CIC2, 1=CH2 ext src mux. CH3 backend input sel 0=CIC3, 1=CH3 ext source mux. FUNCTION
1 0
19:17 16 15:14 13 12 11 10 9 8
42
HSP50216
TABLE 36. BUS ROUTING CONTROL REGISTER (GWA = F801h) (Continued) P(31:0) 7 6 5 4 3 2 1 0 FUNCTION CH0 Ext AGC input enable. 0=CH0 loop filt, 1=external input. CH1 Ext AGC input enable 0=CH1 loop filt, 1=external input. CH2 Ext AGC input enable 0=CH2 loop filt, 1=external input. CH3 Ext AGC input enable Set to 0. CH0 enable serial output 1=FIR0 out enabled to serial outputs. CH1 enable serial output 1=FIR1 out enabled to serial outputs. CH2 enable serial output 1=FIR2 out enabled to serial outputs. CH3 enable serial output 1=FIR3 out enabled to serial outputs. TABLE 37. RESET/SYNC/INTERRUPT SOURCE SELECTION REGISTER (GWA = F802h) P(31:0) 31 30 FUNCTION When set, an interrupt will be generated on each data output of channel 0 to the output block. Typically, this bit will only be set for one channel. When set, the data input to the part will be disabled (the input enable will be zeroed and held at zero) on a P reset (this is always true for the reset pin, whether this bit is set or not, and additionally, the reset pin sets the input mode to gated). The input enable will be released for the input sample that aligns with the SYNCI signal. This is a method for starting up the processing synchronous with a particular data sample. When this bit is set, the carrier center frequency will be updated from the holding register (IWA = *005h) to the active register on the SYNCI signal. If the bit is set in register IWA = *004h to clear the phase accumulator feedback on loading, this function will synchronize the phase of multiple channels. After initial synchronization, the bit in IWA = *004h can be cleared and updates will be synchronous and phase continuous across channels. When this bit is set, the FIR filter compute engine is reset on SYNCI. Resetting the FIR filter compute engine requires 32 clock (CLK) cycles to initialize the read and write pointers. When this bit is set, the AGC is reset on SYNCI. This bit has the same function as bit 29, but for the timing (resampler) NCO. The bit to zero the phase accumulator feedback is in register IWA = *00Ah. When this bit is set, the CIC decimation counter is reset on SYNCI. When this bit is set, the serial output block is reset on SYNCI. If bit 4 in location GWA F803h is set, the serial clock divider is also reset. Same functions as 31:24 for channel 1. Same functions as 31:24 for channel 2. Same functions as 31:24 for channel 3. TABLE 38. SERIAL CLOCK CONTROL REGISTER (GWA = F803h) P(15:0) 5 4 3 FUNCTION When set to 1, this bit will keep the serial clock disabled after a hardware reset until receipt of the first SYNCI signal. Enables resetting serial clock divider on SYNCI. When enabled, a SYNCI enabled for any of the four serial data outputs in the Reset/Sync register (GWA = F802h, bits 24, 16, 8 or 0) will reset the serial clock divider. SCLK polarity. 1 0 Clock low to high transition occurs at the center of the data bit. Clock high to low transition at the center of the data bit.
29
28 27 26 25 24 23:16 15:8 7:0
43
HSP50216
TABLE 38. SERIAL CLOCK CONTROL REGISTER (GWA = F803h) (Continued) P(15:0) 2:0 SCLK rate. 000 001 010 011 100 101 Serial clock disabled. Serial clock rate is Input CLK Rate. Serial clock rate is Input CLK Rate/2. Serial clock rate is Input CLK Rate/4. Serial clock rate is Input CLK Rate/8. Serial clock rate is Input CLK Rate/16. FUNCTION
Other codes are undefined. TABLE 39. INPUT LEVEL DETECTOR SOURCE SELECT/FORMAT REGISTER (GWA = F804h) P(15:0) 15:13 FUNCTION Channel Input Source Selection. Selects as the data input for the level detector either A(15:0), B(15:0), C(15:0), D(15:0) or the P Test Input register as shown below. 15:13 000 001 010 011 100 Source Selected A(15:0) B(15:0) C(15:0) D(15:0) P Test input register. This is provided for testing and to zero the input data bus when a channel is not in use. The Global Write Address register for the P Test input register is F807h.
12
P Register input enable select 1 = bit 11, 0 = one clock wide pulse on each write to location F808h. Select 0 to write data test data into the part. Select 1 to input a constant or to disable the input for minimum power dissipation when the input level detector section is unused. P input enable. When bit 12 is set, this bit is the input enable for the P register input. Active low. 0=enabled, 1=disabled. Parallel Data Input Format 0 1 Two's complement Offset binary
11 10
9
Fixed/Floating point 0 1 Fixed point Floating point. The 16-bit input bus is divided into mantissa and exponent bits grouped either 13/3 or 14/2 depending on bits 8 and 7. See text.
8:7
Floating point mantissa size select. The 16-bit data input is grouped as a 13/3 or 14/2 mantissa/exponent word. These control bits select the mantissa/exponent grouping, add an offset to the exponent and set the shift control saturation level. 00 01 10 11 11/3 bits 15:5 mantissa, 2:0 exponent 12/3 bits 15:4 mantissa, 2:0 exponent 13/3 bits 15:3 mantissa, 2:0 exponent 14/2 bits 15:2 mantissa, 1:0 exponent
6:4
De-multiplex control. These control bits are provided to demultiplex an input data stream comprised of a set of multiplexed data streams. Up to 8 multiplexed data streams can be demultiplexed. These control bits select how many clocks after the ENIx signal to wait before taking the input sample. ENIx should be asserted for one clock period and aligned with the first channel of the multiplexed data set. For example, if four streams are multiplexed at half the clock rate, ENIx would align with the first clock period of the first stream, the second would start two clocks later, the next 4 clocks after ENIx, etc. The samples are aligned with ENIx (zero delay) at the input of the input level detector at the next ENIx. 000 111 zero delay 7 clock periods of delay.
3
Interpolated/Gated Mode Select 0 1 Gated. The input level detector is updated once per clock when ENIx is asserted. Interpolated. The input level detector is updated every clock. The input is zeroed when ENIx is high.
44
HSP50216
TABLE 39. INPUT LEVEL DETECTOR SOURCE SELECT/FORMAT REGISTER (GWA = F804h) (Continued) P(15:0) 2:0 Unused. Set to 0. TABLE 40. INPUT LEVEL DETECTOR CONFIGURATION REGISTER (GWA = F805h) P(31:0) 31:22 21 Set to zero. 1 0 20 1 0 Ones complement of 16-bit data after formatting. Unmodified input. Free run (ignore interval counter). Stop when interval counter times out. FUNCTION FUNCTION
This bit may also be set low temporarily when free running to stabilize the accumulator data for reading. 19:18 Input Level Detector Leak factor, A. 00 01 10 11 17:16 1 2-8 2-12 2-16
Input Level Detector Mode 00 01 10 Leaky integrator (Yn = A*Xn + (1-A)*Yn-1, where A is the gain selected in bits 19:18). Peak detector. Integrator (bit 20 should be set to 0).
15:0
Input Level Detector Interval Load with two less than the desired number of input samples. The interval range is 2 to 65537 input samples. TABLE 41. INPUT LEVEL DETECTOR START STROBE REGISTER (GWA = F806h)
P(15:0) N/A
FUNCTION Writing to this location clears the input level detector accumulator and restarts the interval counter. When the interval counter is done, bit 1 of the status word is set. TABLE 42. P/TEST INPUT BUS REGISTER (GWA = F807h)
P(15:0) 15:0
FUNCTION This 16-bit value can be used as the input to one or more NCO/Mixer/CIC sections or to the input level detector for test or to set the input to a constant value to minimize power when the channel is not in use. The ENI signal for this input is either bit 11 in the channel register at IWA *000h or the strobe generated by a write to location GWA F808h (selected via bit 12 of the channel register at IWA *000h). TABLE 43. P/TEST INPUT BUS ENI REGISTER (GWA = F808h)
P(15:0) N/A
FUNCTION A write to this location, generates and ENI strobe for the P driven input port (when selected via bit 12 of IWA *000h). TABLE 44. SYNCO STROBE REGISTER (GWA = F809h)
P(15:0) N/A
FUNCTION A write to this location will cause a one-clock-wide pulse on the SYNCO pin. The SYNCO pin is used to synchronize multiple channels or parts. The SYNCO pin from one part is typically connected to the SYNCI pin of all the parts. Up to two pipeline registers may be inserted in the SYNCO to SYNCI path.
45
HSP50216
TABLE 45. P FIFO READ ORDER CONTROL REGISTER (GWA = F820h THROUGH F83Fh) P(15:0) 4:0 FUNCTION The five bits selecting the data type are encoded as follows: C C D D D, where CC is the channel number and DDD is the data type. DDD 000 001 010 011 100 101 110 111 I(23:8) I(7:0),8*zeros Q(23:8) Q(7:0),8*zero Mag(23:8) Mag(7:0),8*zero Phase(15:0) AGC gain (15:0) Data Type The upper 16 bits of the I data path via the FIFO/AGC. The lower 8 bits of the I data path. The upper 16 bits of the Q data path via the FIFO/AGC. The lower 8 bits of the Q data path. The upper 16 bits of magnitude (after the gain adjust described in channel register) The lower 8 bits of magnitude. The upper 16 bits of phase. The upper 16 bits of the AGC gain.
Table of Indirect Read Address (IRA) Registers
The address decoding for the read source locations is given below. The internal address of the data to be read is written to direct address 3 (ADD(2:0) = 3) to select and/or fetch the data. A strobe is generated, if needed, to fetch or stabilize the data for reading. If a strobe is needed, the indirect read address must be written to direct address 3 each time the data is needed. If a strobe is not needed, the data can be read repeatedly at direct addresses 0 and 1(ADD(2:0) = 0 and 1, respectively) with any changes in the data showing up immediately. The strobe to sample the AGC gain is generated separately by an indirect write (see IWA *00Fh in the Tables of Indirect Write Address (IWA) Registers). This allows the AGC gain of all the channels to be sampled simultaneously.
NOTE: These Indirect Read Addresses are repeated for each channel. In the addresses below, the * field is the channel select nibble. These bits of the Indirect Address select the target channel register for the data being read. Values of 0 through 3 and F are valid. TABLE 46. TABLE OF INDIRECT READ ADDRESS (IRA) REGISTERS IRA *006h *00Ch *009h *00Fh *100h - *17Fh *180h - *1FCh *400h - *43Fh *440h - *47Fh *480h - *4FFh *500h - *5FFh F806h Active Carrier NCO Center Frequency. Wait Preload, Decr 1&2. Active Timing NCO Center Freq (Most Significant 32 bits). AGC gain (must first write to AGC gain read strobe register IWA = *00Fh before reading). Instruction RAMs. Instruction RAMs (pointer DRAM). Coefficient ROM -HBF, const. Coefficient RAM -1. Coefficient RAM -2. Coefficient ROM -Resampler. Input Level Detector Output. FUNCTION
46
HSP50216
Absolute Maximum Ratings
Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .+6V Input, Output or I/O Voltage . . . . . . . . . . . . GND -0.5V to VCC +0.5V ESD Classification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Class III
Thermal Information
Thermal Resistance (Typical)
JA (C/W)
Operating Conditions
Voltage Range . . . . . . . . . . . . . . . . . . . . . . . . . . . . +3.15V to +3.45V Temperature Range Industrial. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-40C to 85C Input Low Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0V to +0.8V Input High Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .2V to VCC
196 Lead BGA Package (Note 5). . . . . . . . . . . . . . . 27 w/200 LFM Air Flow . . . . . . . . . . . . . . . . . . . . . . . . . 24 w/400 LFM Air Flow . . . . . . . . . . . . . . . . . . . . . . . . . 23 Maximum Junction Temperature . . . . . . . . . . . . . . . . . . . . . . . 150C Maximum Storage Temperature Range . . . . . . . . . . . -65C to 150C Maximum Lead Temperature (Soldering 10s) . . . . . . . . . . . . . 300C
CAUTION: Stresses above those listed in "Absolute Maximum Ratings" may cause permanent damage to the device. This is a stress only rating and operation of the device at these or any other conditions above those indicated in the operational sections of this specification is not implied.
NOTE: 5. JA is measured with the component mounted on a high effective thermal conductivity test board in free air. See Tech Brief TB379 for details.
Electrical Specifications
PARAMETER Logical One Input Voltage Logical Zero Input Voltage Output High Voltage Output Low Voltage Input Leakage Current Output Leakage Current Standby Power Supply Current Operating Power Supply Current Input Capacitance Output Capacitance NOTES:
VCC = 3.3V 0.15V, TA = -40C to 85C, Industrial SYMBOL VIH VIL VOH VOL II IO ICCSB ICCOP CIN COUT TEST CONDITIONS VCC = 3.45V VCC = 3.15V IOH = -2mA, VCC = 3.15V IOL = 2mA, VCC = 3.15V VIN = VCC or GND, VCC = 3.45V VIN = VCC or GND, VCC = 3.45V VCC = 3.45V, Outputs Not Loaded, No CLK f = 70MHz, VIN = VCC or GND, VCC = 3.45V, Outputs Not Loaded Freq = 1MHz, VCC open, all measurements are referenced to device ground MIN 2.0 2.6 -10 -10 MAX 0.8 0.4 10 10 500 850 7 7 UNITS V V V V A A A mA (Note 6) pF (Note 7) pF (Note 7)
6. Power Supply current is proportional to frequency of operation and programmed configuration of the part. Typical rating for ICCOP is 11mA/MHz. 7. Capacitance: TA = 25C, controlled via design or process parameters and not directly tested. Characterized upon initial design and at major process or design changes.
Electrical Specifications
INPUT AND CONTROL TIMING CLK Frequency CLK High CLK Low
VCC = 3.3V 0.15V, TA = -40C to 85C Industrial PARAMETER SYMBOL MIN MAX UNITS
fCLK tCH tCL tDS tDH tPDC tRW tRS
5 5 6 0 5 6
70 6.5 -
MHz ns ns ns ns ns ns ns
Setup Time - Data Inputs, Input Enables, SYNCI to CLK High Hold Time - Data Inputs, Input Enables, SYNCI to CLK High CLK to Output Valid - SYNCO, INTRPT RESET Pulse Width Low RESET Setup Time to CLK High (Note 8)
47
HSP50216
Electrical Specifications
Output Rise, Fall Time (Note 9) MICROPROCESSOR WRITE TIMING P(15:0) Setup Time to Rising Edge of WR P(15:0) Hold Time from Rising Edge of WR A(1:0) Setup Time to Rising Edge of WR A(1:0) Hold Time from Rising Edge of WR CE Setup Time to Rising Edge of WR CE Hold Time from Rising Edge of WR WR Low Time MICROPROCESSOR READ TIMING A(1:0) Setup Time to FALLING Edge of RD A(1:0) Hold Time from RISING Edge of RD RD Enable Time RD Disable Time (Note 9) RD to P(15:0) Data Valid Time CE Setup Time to Falling Edge of RD CE Hold Time from Rising Edge of RD SERIAL CLOCK OUTPUT TIMING CLK to Serial Data, Sync and SCLK (Divide-by 2 through 16 Modes) CLK Low to SCLK Low (Divide-by 1 Mode, Note 9) CLK High to SCLK High (Divide-by 1 Mode, Note 9) Time Skew Between SCLK and Serial Data or Serial Sync (Divide-by 2 through 16 Modes, Note 9) Time Skew Between SCLK and Serial Data or Serial Sync (Divide-by 1 Mode, Note 9) NOTES: 8. The HSP50216 goes into reset immediately on RESET going low and comes out of reset on the 4th rising edge of CLK after RESET goes high. 9. Controlled via design or process parameters and not directly tested. Characterized upon initial design and at major process or design changes. tPD tPDL tPDH tSKEW1 tSKEW2 -1 0.5 6.5 6.5 3 1 2 ns ns ns ns ns tASR tAHR tRE tRD tDV tCSR tCHR 8 -2 8 -2 11.5 8 12 ns ns ns ns ns ns ns tDSW tDHW tASW tAHW tCSW tCHW tWL 10 -2 10 -2 10 -2 5 ns ns ns ns ns ns ns VCC = 3.3V 0.15V, TA = -40C to 85C Industrial (Continued) PARAMETER SYMBOL tRF MIN MAX 3 UNITS ns
AC Test Load Circuit
DUT
S1
CL (NOTE)
NOTE - TEST HEAD CAPACITANCE, 40pF (TYP) SWITCH S1 OPEN FOR ICCSB AND ICCOP EQUIVALENT CIRCUIT IOH 1.5V IOL
48
HSP50216 Waveforms
1/fCLK
tCH
tCL
CLK tDS AIN, BIN, CIN, DIN, ENIA, ENIB, ENIC, ENID, SYNCI tPDC tDH
SYNCO, INTRPT tRW tRS
RESET
FIGURE 3. INPUT AND CONTROL TIMING
RD
CE
WR
ADD(1:0) P(15:0)
tDSW tASW tWL
tDHW tAHW
tCSW
tCHW
FIGURE 4. MICROPROCESSOR WRITE TIMING
49
HSP50216 Waveforms
(Continued)
RD
CE
WR
ADD(1:0)
P(15:0)
tRE
tDV tASR tCSR
tRD tAHR
tCHR
FIGURE 5. MICROPROCESSOR READ TIMING
CLK
SCLK (/2 THROUGH /16)
SCLK (DIVIDE BY 1) tPDH tPDL
SYNC tSKEW
SDXX
tPD
FIGURE 6. SERIAL OUTPUT TIMING
2.0V 0.5V
tRF
tRF
FIGURE 7. OUTPUT RISE AND FALL TIMES
50
HSP50216 ROMd FIR Filters - Response Curves
0.0 -1.0 -2.0 N=1 dB dB -3.0 -4.0 -5.0 N=4 -6.0 0.0 0.1 0.2 fS/R 0.3 0.4 0.5 N=2 N=5 N=3 -60 -80 -100 -120 -140 0.00 N=2 N=3 N=4 N=5 0.10 0.20 fS/R 0.30 0.40 0.50 0 -20 -40 N=1
FIGURE 8. CIC PASSBAND ROLLOFF (N = # OF STAGES, R = DECIMATION FACTOR, fS/R = 1 is CIC OUTPUT RATE)
0
FIGURE 9. CIC FIRST ALIAS LEVEL (N = # OF STAGES, R = DECIMATION FACTOR, fS/R = 1 is CIC OUTPUT RATE)
0 -20 -40 -60 dB -80 -100 -120 -140 0.0 0.5 1.0 1.5 fS/R 2.0 2.5 3.0 -10 -20 -30 -40 -50 -60 -70 -80 -90 -100 -110 -120 0 0.125 HBF2 HBF1
dB
HBF5 HBF4 HBF3
0.25 fS
0.375
0.5
NOTE: HBF4 not included in the ROMd Fir Filter Coefficient memory. See Note 10 of Table 48. FIGURE 10. 5TH ORDER (N = 5) CIC RESPONSE (R = DECIMATION FACTOR, fS/R = 1 is CIC OUTPUT RATE) FIGURE 11. ROMd HALFBAND FILTER FREQUENCY RESPONSE
0 -10 -20 -30 -40 -50 -60 -70 -80 -90 -100 -110 -120 0
HBF3 HBF2 HBF1 HBF5 HBF4
dB
0.0625
0.125 fS
0.1875
0.25
NOTE: HBF4 not included in the ROMd Fir Filter Coefficient memory. See Note 10 of Table 48. FIGURE 12. ROMd HALFBAND FILTER ALIAS FREQUENCY RESPONSE
51
HSP50216 ROMd FIR Filters - Response Curves
(Continued)
10 0 -20 MAGNITUDE (dB) -40 -60 -80 MAGNITUDE (dB) 0 -10 -20 -30 -40 -50 -60 -70 -80 0.5 0.25 0.125 0.375 0.625 0.75 0.0625 0.1875 0.3125 0.4375 0.5625 0.6875 0.8125 0.875 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16 0 FREQUENCY (RELATIVE TO fS) 0.9375 1
-100 -120
FREQUENCY (RELATIVE TO fS)
NOTE: There is a 65dB limitation in SNR using the Re-Sampler Filter. FIGURE 13. POLYPHASE RESAMPLER FILTER BROADBAND FREQUENCY RESPONSE FIGURE 14. POLYPHASE RESAMPLER FILTER PASS BAND FREQUENCY RESPONSE
2 1 0 -1 MAGNITUDE (dB) -2 -3 -4 -5 -6 -7 -8 -9 -10 0.125 0.25 0.0625 0.1875 0.3125 0.375 0.4375 0.5 0
FREQUENCY (RELATIVE TO fS)
FIGURE 15. POLYPHASE RESAMPLER FILTER EXPANDED RESOLUTION PASSBAND FREQUENCY RESPONSE
52
HSP50216
TABLE 47. CIC PASSBAND AND ALIAS LEVELS FREQUENCY fS / R 0 0.01 0.02 0.03 0.04 0.05 0.06 0.07 0.08 0.09 0.10 0.11 0.12 0.13 0.14 0.15 0.16 0.17 0.18 0.19 0.20 0.21 0.22 0.23 0.24 0.25 0.26 0.27 0.28 0.29 0.30 0.31 0.32 0.33 0.34 0.35 0.36 0.37 0.38 0.39 0.40 0.41 0.42 0.43 0.44 0.45 0.46 0.47 0.48 0.49 0.50 5TH ORDER PASSBAND 0 -0.007 -0.029 -0.064 -0.114 -0.179 -0.257 -0.351 -0.458 -0.580 -0.717 -0.868 -1.034 -1.214 -1.409 -1.619 -1.844 -2.084 -2.340 -2.610 -2.896 -3.197 -3.514 -3.847 -4.195 -4.560 -4.941 -5.338 -5.752 -6.183 -6.631 -7.096 -7.578 -8.078 -8.596 -9.133 -9.688 -10.262 -10.854 -11.467 -12.099 -12.752 -13.425 -14.119 -14.835 -15.573 -16.333 -17.116 -17.923 -18.754 -19.610 ALIAS <-200 -199.564 -169.041 -151.023 -138.129 -128.048 -119.749 -112.683 -106.522 -101.054 -96.135 -91.662 -87.558 -83.766 -80.241 -76.947 -73.855 -70.943 -68.189 -65.579 -63.098 -60.734 -58.477 -56.319 -54.252 -52.269 -50.363 -48.531 -46.767 -45.066 -43.426 -41.842 -40.311 -38.832 -37.401 -36.015 -34.674 -33.374 -32.114 -30.892 -29.707 -28.557 -27.442 -26.359 -25.308 -24.287 -23.296 -22.334 -21.399 -20.492 -19.610 4TH ORDER PASSBAND 0 -0.006 -0.023 -0.051 -0.091 -0.143 -0.206 -0.280 -0.367 -0.464 -0.573 -0.694 -0.827 -0.971 -1.127 -1.295 -1.475 -1.667 -1.872 -2.088 -2.317 -2.558 -2.811 -3.077 -3.356 -3.648 -3.953 -4.271 -4.602 -4.946 -5.305 -5.677 -6.063 -6.463 -6.877 -7.306 -7.750 -8.209 -8.684 -9.174 -9.679 -10.201 -10.740 -11.295 -11.868 -12.458 -13.066 -13.693 -14.339 -15.003 -15.688 ALIAS <-200 -159.651 -135.233 -120.818 -110.503 -102.438 -95.799 -90.146 -85.218 -80.843 -76.908 -73.330 -70.047 -67.013 -64.193 -61.558 -59.084 -56.754 -54.551 -52.463 -50.478 -48.587 -46.782 -45.055 -43.402 -41.815 -40.291 -38.825 -37.413 -36.053 -34.740 -33.473 -32.249 -31.066 -29.921 -28.812 -27.739 -26.699 -25.691 -24.713 -23.766 -22.846 -21.953 -21.087 -20.246 -19.430 -18.637 -17.867 -17.119 -16.393 -15.688 3RD ORDER PASSBAND 0 -0.004 -0.017 -0.039 -0.069 -0.107 -0.154 -0.210 -0.275 -0.348 -0.430 -0.521 -0.620 -0.728 -0.846 -0.972 -1.107 -1.251 -1.404 -1.566 -1.737 -1.918 -2.108 -2.308 -2.517 -2.736 -2.965 -3.203 -3.451 -3.710 -3.978 -4.257 -4.547 -4.847 -5.158 -5.480 -5.813 -6.157 -6.513 -6.880 -7.260 -7.651 -8.055 -8.472 -8.901 -9.344 -9.800 -10.270 -10.754 -11.253 -11.766 ALIAS <-200 -119.738 -101.425 -90.614 -82.877 -76.829 -71.849 -67.610 -63.913 -60.633 -57.681 -54.997 -52.535 -50.260 -48.145 -46.168 -44.313 -42.566 -40.913 -39.347 -37.859 -36.440 -35.086 -33.792 -32.551 -31.361 -30.218 -29.119 -28.060 -27.040 -26.055 -25.105 -24.187 -23.299 -22.440 -21.609 -20.804 -20.024 -19.268 -18.535 -17.824 -17.134 -16.465 -15.815 -15.185 -14.572 -13.978 -13.400 -12.840 -12.295 -11.766 2ND ORDER PASSBAND 0 -0.003 -0.011 -0.026 -0.046 -0.071 -0.103 -0.140 -0.183 -0.232 -0.287 -0.347 -0.413 -0.486 -0.564 -0.648 -0.738 -0.834 -0.936 -1.044 -1.158 -1.279 -1.406 -1.539 -1.678 -1.824 -1.976 -2.135 -2.301 -2.473 -2.652 -2.838 -3.031 -3.231 -3.439 -3.653 -3.875 -4.105 -4.342 -4.587 -4.840 -5.101 -5.370 -5.648 -5.934 -6.229 -6.533 -6.847 -7.169 -7.502 -7.844 ALIAS <-200 -79.825 -67.617 -60.409 -55.252 -51.219 -47.900 -45.073 -42.609 -40.422 -38.454 -36.665 -35.023 -33.507 -32.096 -30.779 -29.542 -28.377 -27.276 -26.231 -25.239 -24.294 -23.391 -22.528 -21.701 -20.907 -20.145 -19.412 -18.707 -18.026 -17.370 -16.737 -16.125 -15.533 -14.960 -14.406 -13.869 -13.349 -12.845 -12.357 -11.883 -11.423 -10.977 -10.544 -10.123 -9.715 -9.318 -8.933 -8.560 -8.197 -7.844 1ST ORDER PASSBAND 0 -0.001 -0.006 -0.013 -0.023 -0.036 -0.051 -0.070 -0.092 -0.116 -0.143 -0.174 -0.207 -0.243 -0.282 -0.324 -0.369 -0.417 -0.468 -0.522 -0.579 -0.639 -0.703 -0.769 -0.839 -0.912 -0.988 -1.068 -1.150 -1.237 -1.326 -1.419 -1.516 -1.616 -1.719 -1.827 -1.938 -2.052 -2.171 -2.293 -2.420 -2.550 -2.685 -2.824 -2.967 -3.115 -3.267 -3.423 -3.585 -3.751 -3.922 ALIAS <-200 -39.913 -33.808 -30.205 -27.626 -25.610 -23.950 -22.537 -21.304 -20.211 -19.227 -18.332 -17.512 -16.753 -16.048 -15.389 -14.771 -14.189 -13.638 -13.116 -12.620 -12.147 -11.695 -11.264 -10.850 -10.454 -10.073 -9.706 -9.353 -9.013 -8.685 -8.368 -8.062 -7.766 -7.480 -7.203 -6.935 -6.675 -6.423 -6.178 -5.941 -5.711 -5.488 -5.272 -5.062 -4.857 -4.659 -4.467 -4.280 -4.098 -3.922
53
HSP50216
TABLE 48. DECIMATING HALFBAND FIR FILTER COEFFICIENTS DECIMATING HALFBAND #1 (DHBF #1, 7-TAP) COEFF C0 C1 C2 C3 C4 C5 C6 C7 C8 C9 C10 C11 C12 C13 C14 C15 C16 C17 C18 C19 C20 C21 C22 NOTES: 10. Decimating Halfband Filter #4 Coefficients are shown for reference only and if it is desired to implement this FIR filter these coefficients would have to be loaded into the FIR Coefficient RAM (They are not included in the ROMd Fir Filter Coefficient memory). 11. The 22-bit ROMd FIR filter coefficients are located in the upper 22 bits of the Read register when read back from ROM memory (except for Halfband #4). These bits occupy the upper six bytes (24 bits) with the two LSBs of the lower byte (bits 9:8 of 31:0) being zero. The decimal value for the hexadecimal coefficient is calculated by first converting the hexadecimal value to decimal and the dividing by 223 (8388608). HEX FBFE40 000000 240100 3FFE80 240100 000000 FBFE40 DECIMAL DECIMATING HALFBAND #2 (DHBF #2, 11-TAP) HEX DECIMAL DECIMATING HALFBAND #3 (DHBF #3, 15-TAP) HEX DECIMAL -0.00130558 DECIMATING HALFBAND #4 (DHBF #4, 19-TAP) HEX 000C68 DECIMAL DECIMATING HALFBAND #5 (DHBF #1, 23-TAP) HEX DECIMAL -0.000347137 0.000000000 0.00251293 0.000000000 -0.010158539 0.000000000 0.03055191 0.000000000 -0.081981659 0.000000000 0.309417725 0.500000000 0.309417725 0.000000000 -0.081981659 0.000000000 0.03055191 0.000000000 -0.010158539 0.000000000 0.00251293 0.000000000 -0.000347137
- 0.031303406 00C250 0.000000000 000000 0.281280518 F9B930 0.499954224 000000 0.281280518 258400 0.000000000 3FFF00 - 0.031303406 258400 000000 F9B930 000000 00C250
0.005929947 FFD538 0.000000000 000000 -0.049036026 0195A8 0.000000000 000000 0.29309082 F83FE0
0.000378609 FFF4A0 0.000000000 000000 -0.003810883 005258 0.000000000 000000 0.019245148 FEB320 0.000000000 000000 -0.069904327 03E920 0.000000000 000000 0.304092407 F581A0 0.500000000 000000 0.304092407 279B00 0.000000000 400000 -0.069904327 279B00 0.000000000 000000 0.019245148 F581A0 0.000000000 000000 -0.003810883 03E920 0.000000000 000000 0.000378609 FEB320 000000 005258 000000 FFF4A0
0.000000000 000000 0.012379646 FF8320 0.000000000 000000 -0.06055069 0276A0
0.499969482 000000 0.29309082 265480
0.000000000 000000 0.299453735 F70D60 0.499954224 000000 0.299453735 26EC80 0.000000000 400000 -0.06055069 26EC80
0.000000000 3FFE80 -0.049036026 265480 0.000000000 000000 0.005929947 F83FE0 000000 0195A8 000000 FFD538
0.000000000 000000 0.012379646 F70D60 0.000000000 000000 -0.00130558 0276A0 000000 FF8320 000000 000C68
54
HSP50216
TABLE 49. INTERPOLATING HALFBAND FIR FILTER COEFFICIENTS INTERPOLATING HALFBAND #2 (IHBF #2, 15-TAP) COEFF C0 C1 C2 C3 C4 C5 C6 C7 C8 C9 C10 C11 C12 C13 C14 C15 C16 C17 C18 C19 C20 C21 C22 NOTE: 12. The 22-bit ROMd FIR filter coefficients are located in the upper 22 bits of the Read register when read back from ROM memory. These bits occupy the upper six bytes (24 bits) with the two LSBs of the lower byte (bits 9:8 of 31:0) being zero. The decimal value for the hexadecimal coefficient is calculated by first converting the hexadecimal value to decimal and the dividing by 223 (8388608). HEX FFAA24 000000 032B60 000000 F07F40 000000 4CAB00 800000 4CAB00 000000 F07F40 000000 032B60 000000 FFAA24 DECIMAL -0.002620220 0.000000000 0.024761200 0.000000000 -0.121116638 0.000000000 0.598968506 1.000000000 0.598968506 0.000000000 -0.121116638 0.000000000 0.024761200 0.000000000 -0.002620220 INTERPOLATING HALFBAND #1 (IHBF #1, 23-TAP) HEX FFE944 000000 00A4B4 000000 FD6640 000000 07D240 000000 EB0340 000000 4F3600 800000 4F3600 000000 EB0340 000000 07D240 000000 FD6640 000000 00A4B4 000000 FFE944 DECIMAL -0.000693798 0.000000000 0.005026340 0.000000000 -0.020317078 0.000000000 0.061103821 0.000000000 -0.163963318 0.000000000 0.618835449 1.000000000 0.618835449 0.000000000 -0.163963318 0.000000000 0.061103821 0.000000000 -0.020317078 0.000000000 0.005026340 0.000000000 -0.000693798
55
HSP50216
TABLE 50. RESAMPLER FIR FILTER COEFFICIENTS COEFF C 0 / 191 C 1 / 190 C 2 / 189 C 3 / 188 C 4 / 187 C 5 / 186 C 6 / 185 C 7 / 184 C 8 / 183 C 9 / 182 C 10 / 181 C 11 / 180 C 12 / 179 C 13 / 178 C 14 / 177 C 15 / 176 C 16 / 175 C 17 / 174 C 18 / 173 C 19 / 172 C 20 / 171 C 21 / 170 C 22 / 169 C 23 / 168 C 24 / 167 C 25 / 166 C 26 / 165 C 27 / 164 C 28 / 163 C 29 / 162 C 30 / 161 C 31 / 160 NOTE: 13. The 22-bit ROMd FIR filter coefficients are located in the upper 22 bits of the Read register when read back from ROM memory. These bits occupy the upper six bytes (24 bits) with the two LSBs of the lower byte (bits 9:8 of 31:0) being zero. The decimal value for the hexadecimal coefficient is calculated by first converting the hexadecimal value to decimal and the dividing by 223 (8388608). HEX 004000 006910 007A90 008C90 009ED0 00B0E0 00C230 00D240 00E090 00ECC0 00F620 00FBC0 00FCB0 00F970 00EFF0 00E050 00C980 00AAD0 0083B0 005370 0019A0 FFD590 FF86F0 FF2D90 FEC930 FE59C0 FDDF80 FD5A60 FCCB00 FC31F0 FB9000 FAE600 DECIMAL 0.001953125 0.003206253 0.003740311 0.004289627 0.004846573 0.005397797 0.005926132 0.006416321 0.006853104 0.007225037 0.007511139 0.007682800 0.007711411 0.007612228 0.007322311 0.006845474 0.006149292 0.005212784 0.004018784 0.002546310 0.000782013 -0.001295090 -0.003694534 -0.006422043 -0.009485245 -0.012886047 -0.016616821 -0.020679474 -0.025054932 -0.029726028 -0.034667969 -0.039855957 COEFF C 32 / 159 C 33 / 158 C 34 / 157 C 35 / 156 C 36 / 155 C 37 / 154 C 38 / 153 C 39 / 152 C 40 / 151 C 41 / 150 C 42 / 149 C 43 / 148 C 44 / 147 C 45 / 146 C 46 / 145 C 47 / 144 C 48 / 143 C 49 / 142 C 50 / 141 C 51 / 140 C 52 / 139 C 53 / 138 C 54 / 137 C 55 / 136 C 56 / 135 C 57 / 134 C 58 / 133 C 59 / 132 C 60 / 131 C 61 / 130 C 62 / 129 C 63 / 128 HEX FA3540 F97F00 F8C4C0 F80880 F74C40 F691C0 F5DB80 F52C00 F48600 F3EC00 F36140 F2E880 F284C0 F23980 F20940 F1F7C0 F20800 F23C80 F298C0 F31F00 F3D280 F4B500 F5C900 F71040 F88C40 FA3E80 FC27C0 FE48C0 00A140 033140 05F7C0 08F400 DECIMAL -0.045249939 -0.050811768 -0.056495667 -0.062240601 -0.067985535 -0.073677063 -0.079238892 -0.084594727 -0.089660645 -0.094360352 -0.098594666 -0.102279663 -0.105323792 -0.107620239 -0.109092712 -0.109626770 -0.109130859 -0.107528687 -0.104713440 -0.100616455 -0.095138550 -0.088226318 -0.079803467 -0.069816589 -0.058219910 -0.044967651 -0.030036926 -0.013404846 0.004920959 0.024940491 0.046623230 0.069946289 COEFF C 64 / 127 C 65 / 126 C 66 / 125 C 67 / 124 C 68 / 123 C 69 / 122 C 70 / 121 C 71 / 120 C 72 / 119 C 73 / 118 C 74 / 117 C 75 / 116 C 76 / 115 C 77 / 114 C 78 / 113 C 79 / 112 C 80 / 111 C 81 / 110 C 82 / 109 C 83 / 108 C 84 / 107 C 85 / 106 C 86 / 105 C 87 / 104 C 88 / 103 C 89 / 102 C 90 / 101 C 91 / 100 C 92 / 99 C 93 / 98 C 94 / 97 C 95 / 96 HEX 0C2400 0F8600 131700 16D400 1ABA00 1EC500 22F100 273A00 2B9900 300A00 348800 390C00 3D9100 420F00 468200 4AE200 4F2A00 535200 575400 5B2B00 5ED000 623E00 656E00 685D00 6B0500 6D6200 6F7000 712C00 729200 73A100 745600 74B200 DECIMAL 0.094848633 0.121276855 0.149139404 0.178344727 0.208801270 0.240386963 0.272979736 0.306457520 0.340606689 0.375305176 0.410400391 0.445678711 0.480987549 0.516082764 0.550842285 0.585021973 0.618469238 0.650939941 0.682250977 0.712249756 0.740722656 0.767517090 0.792419434 0.815338135 0.836090088 0.854553223 0.870605469 0.884155273 0.895080566 0.903350830 0.908874512 0.911682129
56
HSP50216
.
TABLE 51. BIT WEIGHTING FOR AGC LOOP FEEDBACK PATH AGC LOOP FILTER GAIN AGC LOOP (EXPONENT) FILTER GAIN AGC LOOP FILTER GAIN MULTIPLIER SHIFT SHIFT SHIFT SHIFT =0 =4 =8 = 15 (OUTPUT) (MANTISSA) 2 2 2 2 15 14 13 12 11 10 9 8 7 6 5 4 3 2 1 0 =2 =1 = 0. =1 =2 =3 =4 =5 =6 =7 =8 =9 = 10 = 11 = 12 = 13 0. x x x x 2 1 0. 1 2 3 4 5 6 7 8 9 10 11 12 13 14 2 2 2 2 2 2 2 2 2 2 2 2 2 2 2 2 1 0. 1 2 3 4 5 6 7 8 9 10 2 2 2 2 2 2 2 2 2 2 2 2 2 2 2 2 1 0. 1 2 3 4 5 6 7 8 9 10 11 12 13 14 2 2 2 2 2 2 2 2 2 2 2 2 1 0. 1 2 3 4 5 6 7 8 9 10 11 12 13 14 G G G G 2 2 2 2 2 1 0. 1 2 3 4 5 6 7 8 9 10 11 12 13 14 G G G G G G G G G G G 3 2 1 0 -1 -2 -3 -4 -5 -6 -7 -8 -9 -10 -11 -12 AGC BIT WEIGHTS TO OUTPUT SECTION 0 E E E E M M M M M M M M M M M M M M TO P 0 E E E E M M M M M M M M M M M 48 24 12 6 3 1.5 0.75 0.375 0.1875 0.09375 0.04688 0.02344 0.01172 0.00586 0.00293 0.00146 0.000732 0.000366 0.000183 0.0000916 0.0000458 0.0000229 0.0000114 0.00000572 0.00000286 AGC GAIN RESOLUTION (dB)
GAIN AGC ERROR GAIN ACCUM BIT ERROR BIT POSITION INPUT WEIGHT 31 30 29 28 27 26 25 24 23 22 21 20 19 18 17 16 15 14 13 12 11 10 9 8 7 6 5 4 3 2 1 0
LIMITS
57
HSP50216 Plastic Ball Grid Array Packages (BGA)
o
A1 CORNER A1 CORNER I.D.
A D
V196.12x12
196 BALL PLASTIC BALL GRID ARRAY PACKAGE INCHES SYMBOL A A1
E
MILLIMETERS MIN 0.31 0.93 0.41 11.90 10.30 MAX 1.50 0.41 1.11 0.51 12.10 10.50 196 0.80 BSC 14 x 14 0.10 0.12 NOTES 7 3 Rev. 2 12/00
MIN 0.012 0.037 0.016 0.468 0.405 196
MAX 0.059 0.016 0.044 0.020 0.476 0.413
A2 b D/E D1/E1
B
N e MD/ME bbb
A1 CORNER A1 CORNER I.D.
TOP VIEW 0.15 MCAB 0.006 0.08 MC 0.003 b
0.032 BSC 14 x 14 0.004 0.005
D1
aaa NOTES:
14 13 12 11 10 9 8 7 6 5 4 3 2 1
S
A
A B C D E F G E1 H J K L M N P e S A BOTTOM VIEW A1 A2 bbb C ALL ROWS AND COLUMNS
1. Controlling dimension: MILLIMETER. Converted inch dimensions are not necessarily exact. 2. Dimensioning and tolerancing conform to ASME Y14.5M-1994. 3. "MD" and "ME" are the maximum ball matrix size for the "D" and "E" dimensions, respectively. 4. "N" is the maximum number of balls for the specific array size. 5. Primary datum C and seating plane are defined by the spherical crowns of the contact balls. 6. Dimension "A" includes standoff height "A1", package body thickness and lid or cap height "A2". 7. Dimension "b" is measured at the maximum ball diameter, parallel to the primary datum C. 8. Pin "A1" is marked on the top and bottom sides adjacent to A1. 9. "S" is measured with respect to datum's A and B and defines the position of the solder balls nearest to package centerlines. When there is an even number of balls in the outer row the value is "S" = e/2.
C A SEATING PLANE SIDE VIEW
aaa C
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58

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